Method and apparatus for wireless spread spectrum communication with preamble sounding gap

ABSTRACT

A wireless spread spectrum communication system comprises a spread spectrum transmitter and spread spectrum receiver which communicate according to an over-the-air protocol. The spread spectrum transmitter transmits a burst comprising a preamble followed by a short gap, followed by a data message. The spread spectrum receiver receives and demodulates the transmitted burst. The receiver detects the preamble using a non-coherent parallel correlator, and from the preamble correlation peak generates a series of integration periods for serial non-coherent correlation. The short gap between the preamble and the data message allows the receiver time to process the preamble and set the timing of the electronics for receiving the data message. The receiver has a plurality of non-coherent correlators operating in parallel to recover the spread spectrum encoded information. For each of M spread spectrum codes, the receiver simultaneously attempts to correlate the non-preamble portion of the received spread spectrum signal by separating the received signal into real and imaginary parts, correlating both real and imaginary parts for I and Q sequences, and combining the real I, real Q, imaginary I, and imaginary Q correlation signals into a unified correlation signal.

This application is a continuation of application Ser. No. 08/928,846,filed on Sep. 12, 1997 now U.S. Pat. No. 5,953,370, which is acontinuation of application Ser. No. 08/480,903, filed on Jun. 7, 1995now abandoned, which is a continuation-in-part of application Ser. No.08/304,091, filed on Sep. 9, 1994, now U.S. Pat. No. 5,648,982.

FIELD OF THE INVENTION

The field of this invention relates to spread spectrum communicationand, more particularly, to transmitting and receiving continuous phasemodulated (CPM) signals such as spread spectrum signals.

DESCRIPTION OF THE RELATED ART

Spread spectrum is a type of signal modulation that spreads a signal tobe transmitted over a bandwidth that substantially exceeds thedata-transfer rate, hence the term “spread spectrum”. In direct sequencespread spectrum, a data signal is modulated with a pseudo-random chipsequence; the encoded spread spectrum signal is transmitted to thereceiver which despreads the signal. Several techniques are availablefor the transmitter to modulate the data signal, including biphase shiftkeying (BPSK) and continuous phase modulated (CPM) techniques. Minimumshift keying (MSK) is a known variation of CPM.

In despreading a spread spectrum signal, the receiver produces acorrelation pulse in response to the received spread spectrum signalwhen the received spread spectrum signal matches the chip sequence to apredetermined degree. Various techniques are available for correlatingthe received signal with the chip sequence, including those usingsurface acoustic wave (SAW) correlators, tapped delay line (TDL)correlators, serial correlators, and others.

In spread spectrum communication CPM techniques are often chosen so asto preserve signal bandwidth of the spread spectrum signal when it isamplified and transmitted. Using CPM techniques also has the advantagethat “class C” amplifiers may be used for transmitting the spreadspectrum signal. However, spread spectrum signals transmitted using CPMare difficult to decode with many types of spread spectrum correlators,including various SAW correlators and serial correlators. These types ofcorrelators usually require a BPSK spread spectrum signal for effectivecorrelation rather than an MSK or other CPM spread spectrum signalbecause a BPSK signal has either a zero or 180 degree phase shift foreach chip time. Thus, each chip of a received BPSK signal may becompared with each chip of the spread spectrum code, and a maximumcorrelation pulse may be generated when a predetermined number ofmatches occur. However, when a CPM signal with the same data signal andchip rate is applied to the same correlator, the correlation pulse willgenerally be very weak and may be quite difficult to detect.

Another problem often encountered in attempting to correlate spreadspectrum signals transmitted using CPM techniques is the absence of acoherent reference signal in the receiver. A coherent reference signalin this sense may be defined as a locally generated signal that matchesthe transmitter carrier signal in frequency and phase. The receiver mayuse the locally generated reference signal to demodulate the receivedsignal. In practice, however, it can be difficult to independentlygenerate a local reference signal in the receiver precisely matching thetransmitted carrier signal in frequency and phase. Rather, a localreference signal generated in the receiver will usually be of anon-coherent variety—that is, having small differences in frequency andphase from the transmitter's carrier signal. These frequency and phasedifferences are not constant but vary over time. When an attempt is madeto demodulate a received signal using a non-coherent reference signal,errors in correlation may occur due to mismatches in timing andvariations in perceived amplitude caused by the frequency and phasedifferences.

Various methods for dealing with the above problem exist in which acoherent reference signal is created in the receiver by continuouslymeasuring the frequency and phase differences between the receivedsignal and a locally generated non-coherent reference signal, and thenadjusting the non-coherent reference signal until it matches thefrequency and phase of the received signal. Such methods, however,generally require the use of relatively complex feedback techniques andinvolve extra hardware. Moreover, locking onto the received frequencyand phase can take an unacceptably large amount of time, particularly insystems where time is of the essence, such as in certain time divisionmultiple access (TDMA) systems in which only a relatively brief timeslot is allocated for periodic communication between a transmitter andreceiver.

A particular non-coherent digital matched filter is described in A.Baier and P. W. Baier, “Digital Matched Filtering of ArbitrarySpread-Spectrum Waveforms Using Correlators with Binary Quantization,” 2Proceedings, 1983 IEEE Military Communications Conference, Vol. 2, pp.418-423 (1983). The digital filter described therein uses four realfilter channels to perform four-phase quantization in the complex plane,with the four quadrants being the quantization regions, and the resulttaking on the four complex values of ±1±j. In the described four-phasefilter, an input signal is divided into an in-phase signal and aquadrature signal. The in-phase signal and the quadrature signal areseparately filtered, sampled and digitized using 1-bit quantization. Thequantized in-phase signal and the quantized quadrature signal are eachfed into two binary correlators each programmed with a referencesequence of N chips, one chip for each sample. The outputs of the fourbinary correlators are combined to produce a resultant output signal.Baier's four-phase digital matched filter is also described in A. Baier,“A Low-Cost Digital Matched Filter for Arbitrary Constant-EnvelopeSpread Spectrum Waveforms,” IEEE Transactions on Communications, Vol.Com-32, No. 4, April 1984, pp. 354-361.

These references suggest that for demodulation of non-coherent CPMsignals such as QPSK, MSK, OQPSK, and GMSK signals, four real channelsare needed to fully recover the transmitted signal. Further, thedescribed four-phase filter shows only a system using 1-bitquantization, and does not describe a technique for serial correlation.

Accordingly, it would be advantageous to provide a method of modulationand demodulation particularly suited to CPM signals. It would further beadvantageous to provide a method of CPM modulation and demodulation thatdoes not require the generation of a coherent reference signal, that iscapable of rapid correlation, and that may be used with analogcorrelators and digital correlators in an effective manner. It wouldfurther be advantageous to provide a flexible and effective system forCPM modulation and demodulation that does not require a coherentreference signal, and that is suitable for use in an environment ofcellular communications.

SUMMARY OF THE INVENTION

The invention relates to a method and apparatus for transmitting andreceiving CPM spread spectrum signals using phase encoding to increasethroughput. In one aspect of the invention, a transmitter divides asignal data stream into a plurality of data streams (e.g., an I and Qdata stream), independently modulates the data streams using CPM or arelated modulation technique, and superposes the plurality of resultantsfor transmission. A preferred receiver receives the superposed spreadspectrum signal, simultaneously attempts to correlate for a plurality ofchip sequences (such as I and Q chip sequences), and interleaves thecorrelated data streams into a unified signal data stream.

In a second aspect of the invention, the receiver comprises a carriersignal that is neither frequency matched or phase matched with thetransmitted signal. In this aspect, the receiver separates the receivedspread spectrum signal into real and imaginary parts, attempts tocorrelate both real and imaginary parts for a plurality of chipsequences (e.g., I and Q chip sequences), and combines the real andimaginary signals into a unified signal data stream. A preferredembodiment of this aspect of the invention uses a single bitdigitization of the received spread spectrum signal to preserve onlyphase information for inexpensive digital processing. Another preferredembodiment of this aspect of the invention uses two-bit digitization ofthe received spread spectrum signal. In an alternative embodiment of theinvention, the receiver uses self-synchronization techniques fordespreading and correlation.

These aspects of the invention are described with reference to apreferred embodiment of the invention, in which a single parallelcorrelator and a plurality of 32 serial correlators are combined so asto allow correlation and recognition of any of 32 distinct symbols for aspread spectrum code sequence of 32 chips. Each of the 32 distinctsymbols is associated with a distinct 5-bit pattern. A sixth bit ofinformation is transmitted for each symbol by differential phaseencoding at the transmitter and is phase decoded at the receiver.

A preferred transmitter capable of phase encoding divides a data streaminto a data symbol portion and a phase selection portion. The datasymbol portion is used to select one of a plurality of symbol codes fortransmission. The phase selection portion is used to differentiallyphase encode the selected symbol code prior to transmission. Thetransmitter may use a CPM or related technique to transmit the phaseencoded symbol codes.

A preferred receiver receives the superposed spread spectrum signal andsimultaneously attempts to correlate for a plurality of chip sequences(such as I and Q chip sequences), and derives a real correlation signaland an imaginary correlation signal. For each received symbol, thereceiver determines which is of a plurality of phase sectors the phaseangle lies in. The receiver compares the difference between the phasesector of the present symbol and the phase sector of a preceding symbol.For biphase encoding, if the difference in closer to 0°, then thereceiver outputs a first bit, and if the difference is closer to 180°,the receiver outputs a second bit. Higher degrees of phase encoding(e.g., quadraphase or octiphase) may also be used.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a spread spectrum communication transmitterand receiver as known in the art.

FIG. 2 depicts a pattern of cells for use in spread spectrumcommunication.

FIG. 3 is a graph of phase changes over time for an MSK signal.

FIGS. 4A-4C are a set of graphs showing a relationship among phasecomponents.

FIG. 5A is a block diagram showing means for generating a CPM spreadspectrum signal.

FIG. 5B is a graph of I and Q values.

FIG. 6 is a block diagram of a spread spectrum transmitter.

FIG. 7 is a block diagram showing one embodiment of spread spectrumreceiver.

FIG. 8 is a block diagram showing another embodiment of a spreadspectrum receiver.

FIG. 9 is a scatter diagram comparing transmitted and received I and Qsignals.

FIG. 10 is a block diagram of an embodiment of a spread spectrumreceiver using separable real and imaginary parts of a received spreadspectrum signal.

FIGS. 11A-11F are diagrams showing a representation of transmitted andreceived waveforms for different phase values.

FIG. 12 is a block diagram of another embodiment of a spread spectrumreceiver using separable real and imaginary parts of a received spreadspectrum signal.

FIG. 13A is a block diagram of an embodiment of a spread spectrumreceiver using serial correlation, and FIG. 13B is a waveform diagramassociated therewith.

FIG. 14 is a block diagram of an embodiment of spread spectrum receiverusing serial correlation for separable real and imaginary parts of thereceived spread spectrum signal.

FIG. 15A is a block diagram of another embodiment of a spread spectrumreceiver using serial correlation for separable real and imaginary partsof the received spread spectrum signal.

FIG. 15B is a block diagram of a spread spectrum receiver usingmulti-bit serial correlation for separable real and imaginary parts ofthe received spread spectrum signal.

FIG. 15C is a graph showing an example of quantization of an I or Qwaveform in accordance with the FIG. 15B receiver.

FIG. 15D is a block diagram of another embodiment of a spread spectrumreceiver using multi-bit serial correlation for separable real andimaginary parts of the received spread spectrum signal.

FIG. 16 is a block diagram of an embodiment of spread spectrum receiverusing self-synchronized correlation for separable real and imaginaryparts of the received spread spectrum signal.

FIGS. 17A and 17D are block diagrams of a preferred transmitter and apreferred transmission protocol, respectively.

FIG. 17B is a diagram of an alternative transmission protocol.

FIG. 17C is an exemplary SQAM waveform generated by a transmitter usingseparate I and Q components.

FIG. 18 is a block diagram of a preferred noncoherent matched filter andassociated receiver components.

FIG. 19 is a block diagram of a preferred digital circuit embodiment ofa set of noncoherent serial correlators and associated receivercomponents.

FIG. 20 is a diagram showing exemplary correlation pulses within apredetermined timing window.

FIGS. 21A and 21B are schematic diagrams showing a preferred digitalcircuit embodiment of part of a receiving system used in conjunctionwith the circuitry of FIGS. 18 and 19.

FIG. 22 is a block diagram of a Robertson device for computing a sum ofthe squares of its inputs.

FIG. 23 is a block diagram of a correlator matched to a specific codesequence.

FIGS. 24A and 24B are digital circuit block diagrams of a spreadspectrum transmitter employing differential phase encoding, and

FIG. 24C is a general block diagram thereof.

FIG. 24D is a diagram of an exemplary input data sequence and phaseencoded symbol code output sequence.

FIGS. 25A and 25B-25C are block diagrams of two different embodiments ofa receiver for carrying out phase decoding to obtain extra informationfrom the received signal.

FIG. 26 is a block diagram of a preferred receiver for carrying outphase decoding in a 32 symbol transmission technique in accordance withthe embodiment of the receiver shown in FIGS. 25B and 25C.

FIGS. 27A and 27B are phase map diagrams for an 8-sector phase map and a16-sector phase map, respectively, and

FIG. 27C is a preferred 16-sector phase map diagram having a phasereference offset from zero.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 is a block diagram of a spread spectrum communication transmitter101 and receiver 108 as known in the art.

The spread spectrum transmitter 101 of FIG. 1 comprises an input port102 for input data 103, a transmitter chip sequence generator 104, and amodulator 105. The transmitter 101 thereby transmits a spread spectrumsignal 106 over a transmission channel 107. The transmission channel 107may comprise an RF channel, but may also comprise other transmissionmedia, such as modulated laser, ultrasound, or fluidic systems. Thespread spectrum receiver 108 of FIG. 1 comprises a receiver chipsequence generator 110, a demodulator 111, and an output port 112 forgenerating output data 113. In the FIG. 1 system, a single chipsequence, which appears essentially random to others not knowing thespreading code upon which it is based, may be identically generated byboth the transmitter generator 104 and the receiver generator 110. Anextensive discussion of spread spectrum communication, spreading codes,and chip sequences may be found in R. Dixon, Spread Spectrum Systemswith Commercial Applications (J. Wiley & Sons, 3d ed. 1994).

FIG. 2 depicts a pattern of cells for use in spread spectrumcommunication.

In the preferred cellular environment of FIG. 2, a region 151 forcommunication may be divided into a set of cells 152, each of which maybe assigned a frequency and a set of spread spectrum codes forcommunication. A first cell 153 may generally be found adjacent to a setof distance-one neighbors 154 and a set of distance-two neighbors 155.In a preferred embodiment, a plurality of frequencies f1, f2 and f3, anda plurality of code sets c1, c2, and c3, may be configured in a patternof cells 152 so that the no distance-one neighbors 154 or distance-twoneighbors 155 of a particular cell 153 has the same combination offrequency and code set as the cell 153.

Other and further information about a preferred cellular environment inwhich the invention may operate may be found in U.S. Pat. No. 5,402,413which is assigned to the assignee of the present application, and herebyincorporated by reference as if fully set forth herein.

Known CPM spread spectrum signals include several variations; theseinclude minimum shift keying (MSK) and its variations, e.g., Gaussianpre-filtered MSK (GMSK), superposed quadrature amplitude modulation(SQAM), and staggered quadrature offset raised cosine modulation(SQORC). These variations are known in the art. Explanations of varioustypes of CPM techniques may be found in the following: Frank Amoroso andJames A. Kivett, “Simplified MSK Signaling Technique,” IEEE Transactionson Communications, April 1977, pp. 433-441; Mark C. Austin and Ming U.Chang, “Quadrature Overlapped Raised-Cosine Modulation,” IEEETransactions on Communications, Vol. Com-29, No. 3, March 1981, pp.237-249; Kazuaki Murota and Kenkichi Hirade, “GMSK Modulation forDigital Mobile Radio Telephony,” IEEE Transactions on communications,Vol. Com-29, No. 7, July 1981, pp. 1044-1050; and J. S. Seo and K.Feher, “SQAM: A New Superposed QAM Modem Technique,” IEEE Transactionson Communications, Vol. Com-33, March 1985, pp. 296-300. The inventionis generally described with regard to MSK signals. However, othervariants of MSK and other CPM signals are within the scope and spirit ofthe invention.

An MSK signal is generally characterized by the fact that phase changeslinearly within each chip time, and that the phase change over a singlechip time is ±π/2 radians (±90 degrees). The rate of phase change for asingle chip time is ±k, for a suitable value k, and is linear andcontinuous everywhere except at chip boundaries.

The above described characteristics of MSK signals may be furtherexplained with reference to FIG. 3, which is a graph showing possiblechanges in phase for an MSK signal over time. In FIG. 3, the x-axis istime and the y-axis is signal phase. In a first chip time from zero toTc, the phase θ(t) changes from 0 to π/2 or −π/2. In a second chip time,from Tc to 2 Tc, the phase θ(t) changes from +π/2 to 0 or +π/2 to +π, orfrom −π/2 to 0 or −π/2 to −π, and so on.

An MSK signal s(t) may be considered to comprise two offset signals,i(t) and q(t), which represent the phase of the carrier signal. At anyinstant of time the phase of the carrier signal may be expressed as:

θ(t)=−Tan⁻¹[q(t)/i(t)]

Thus, i(t)=cos θ(t), and q(t)=sin θ(t).

Since the phase of the MSK signal varies linearly from one chip time tothe next chip time, i(t) and q(t) may consist of half sinusoidalwaveforms as shown in the FIGS. 4A-4C. In FIGS. 4A-4C, the x-axis istime and the y-axis is signal phase. FIG. 4A is a graph showing anexample of how the phase θ(t) may change for a particular MSK signal ineach chip time from 0, Tc, 2Tc, 3Tc, 4Tc, 5Tc, and so on, for the chipsequence “11101001 . . . .” As noted, during each chip time the phasevaries for an MSK signal by π/2 in either a positive or negativedirection. FIGS. 4B and 4C are graphs showing i(t) and q(t) waveforms,respectively, which correspond to the varying phase θ(t). Because of thenature of the MSK signal's phase θ(t) (i.e., that it is linear andvaries only by π/2 each chip period), the i(t) signal comprises asequence of partial cosine waveforms, and the q(t) signal comprises asequence of partial sine waveforms. Each of i(t) and q(t) comprises ahalf-waveform over a timespan of 2Tc; that is, i(t) and q(t) occur athalf the chip rate.

An i(t) waveform and a q(t) waveform can be generated from a chip streamc(t) and combined so as to produce an MSK signal—i.e., a signal having aphase which varies linearly as desired in either a positive or anegative direction by an amount of π/2 each chip time. In order togenerate i(t) and q(t), the original chip stream c(t) may bedemultiplexed into two separate chip streams C_(even)(t) and C_(odd)(t),each having half the chip rate of the original chip stream c(t). In thedescribed embodiment, the i(t) signal is associated with theodd-numbered chips, and the q(t) signal is associated with theeven-numbered chips.

Thus, the i(t) signal comprises a sequence of half-sinusoidal waveforms,one for each odd chip. Each half sinusoid may be positive for a “1” chipand negative for a “0” chip:

i(t)=C_(odd)(t)cos θ(t)  (203)

where C_(odd)(t) comprises the odd-numbered chips from the chip streamto be transmitted. Similarly, the q(t) signal comprises a sequence ofhalf-sinusoidal waveforms, one for each even chip:

q(t)=C_(even)(t)sin θ(t)  (204)

where C_(even)(t) comprises the even-numbered chips from the chip streamto be transmitted.

The i(t) and q(t) signals may be used to modulate a carrier signaloperating at frequency ω₀ by summing i(t) and q(t) in phase quadratureso as to generate an MSK signal s(t) having a linearly varying phaseθ(t). A block diagram showing means for generating a CPM spread spectrumsignal is depicted in FIG. 5A. The signal i(t) is multiplied with asignal A cos ω₀t by multiplier 250, which provides an output to a summer252. The signal q(t) is multiplied with a signal A sin ω₀t by multiplier251, which also provides an output to the summer 252. The summer 252sums its inputs and produces an output signal s(t).

The relationship between the transmitted signal s(t) having varyingphase θ(t), and the i(t) and q(t) signals is shown in the followingequations:

s(t) = Re {A exp(j [−ω₀t + θ(t)])} = Re {A exp(−jω₀t) exp(jθ(t))} = Re{A [cos ω₀t − j sin ω₀t] [i(t) + jq(t)]} = A i(t) cos ω₀t + A q(t) sinω₀t (207)

where A is an amplification factor, Re{} represents the real part of acomplex value, and j is the square root of −1. Note that u(t)=i(t)+jq(t)represents the complex envelope of s(t).

As noted herein, i(t) and q(t) each comprises every other chip from thechip stream c(t); i(t) comprises the odd-numbered chips 1, 3, 5, . . . ;q(t) comprises the even-numbered chips 2, 4, 6, . . . . The transmittedsignal s(t), generated from signals i(t) and q(t), therefore comprisesall of the chips. Because q(t) is derived from the even chips while i(t)is derived from the odd chips, q(t) is delayed by one chip time fromi(t); thus, q(t) and i(t) are offset signals.

It is important to note that, because i(t) and q(t) are staggered, asi(t) reaches its maximum (or minimum) value q(t) will be zero, and viceversa. This relationship between i(t) and q(t) allows phase changesequences of ±π/2 over one chip time Tc (unlike, for example, QPSK orOQPSK). FIG. 5B is a graph of I and Q values, in which the x-axisrepresents values of i(t) and the y-axis represents values of q(t). Each<i(t), q(t)> pair falls at a given instant of time on the circle 260.Maximum and minimum values for i(t) and q(t) are shown where the circle260 intersects the x-axis and y-axis at points 265 through 268; thesepoints 265 through 268 also represent the possible values of <i(t),q(t)> pairs at chip boundary times.

Alternative encoding methods such as GMSK, SQAM, or SQORC, differ fromMSK in that phase changes of less than ±π/2 are allowed. In general,GMSK, SQAM, and SQORC all use a form of pre-filtering the MSK i(t) andq(t) signals to reduce transmission bandwidth. This pre-filtering hasthe general effect of reducing the high-frequency components generatedby the sharp phase reversals in the MSK i(t) and q(t) signals. For GMSK,pre-filtering may also result in intersymbol interference over severalchip times, the effect of which may be mitigated with a trellis decoder.In SQAM or SQORC, the final frequency envelope is no longer constant,but is still nearly so.

FIG. 6 is a block diagram of a spread spectrum transmitter.

In the transmitter of FIG. 6, a chip stream c(t) 301 is provided to ademultiplexer 302, which divides the chip stream 301 into a set of oddchips C_(odd)(t) 303 for the i(t) signal and a set of even chipsC_(even)(t) 304 for the q(t) signal. The chip stream c(t) 301 maycomprise the result of a pseudo-noise (“PN”) code modulated with a datastream (as in direct sequence spread spectrum communication), or maycomprise a sequence of chip codes corresponding to predetermined symbolssuch as may be done, for example, in code shift keying (CSK) techniques.

The odd chips 303 and the even chips 304 are each coupled to first andsecond waveform generators P(t) 305 and 306 respectively. In a preferredembodiment, the waveform generators P(t) may generate a half-sinusoidalwaveform, positive or negative, as described herein. Other waveformgenerators and other waveforms are within the scope and spirit of theinvention.

The output of the first waveform generator 305 (i.e., receiving the oddchips 303) corresponds to the signal i(t) and is coupled to a firstmultiplier 307, which modulates a carrier signal cos ω₀t to generate asignal s₁(t) 308 corresponding to i(t) cos ω₀t. The output of the secondwaveform generator 306 (i.e., receiving the even chips 304) correspondsto the signal q(t), which, as mentioned, is delayed by one chip time Tcfrom the signal i(t). The output of the second waveform generator 306 iscoupled to a second multiplier 310, which modulates a carrier signal sinw₀t to generate a signal s₂(t) 311 corresponding to q(t) sin w₀t.

The signals s₁(t) 308 and s₂(t) 311 are coupled to a summer 312, whichcombines its inputs and generates a superposed signal s(t) 313. Thesignal s(t) may be amplified and transmitted by a transmission system,such as a radio transmission system, coupled to the transmission channel107.

The chip stream c(t) may be generated by modulating a pseudo-noise codewith data to be transmitted such as is known in direct sequence spreadspectrum modulation. In a preferred embodiment, the chip stream c(t)comprises a plurality of symbol codes, each symbol code representing asymbol indicative of one or more data bits of information. Instead ofdirectly modulating input data with a pseudo-noise code, sequences ofdata bits are translated into symbols which are used to select from aplurality of symbol codes located in a table. For example, five databits may represent a symbol; thus, there may be 32 possible symbolsrepresenting all possible combinations of five data bits. Each symbol isassociated with a unique symbol code, so that thirty-two symbol codes(or sixteen symbol codes and their inverses) may represent all possiblesymbols. For each symbol to be transmitted, the appropriate symbol codeis selected among the thirty-two available. Thus, the chip stream c(t)may comprise a sequence of symbol codes.

Each symbol code may be, for example, 32 chips in length, or some otherappropriate number of chips in length (preferably an even number ofchips).

In a like manner, the demultiplexer 302 may comprise a table of halfsymbol codes. In particular, the demultiplexer 302 may comprise aQ-lookup table and I-lookup table. For every five bits of data to betransmitted (following the previous example), instead of looking up asymbol code from a table and demultiplexing it with demultiplexer 302,two half symbol codes may be read, one from the I-lookup table and onefrom the Q-lookup table. Each half symbol code may be clocked seriallyto the waveform generators 305, 306 for further processing. The systemmay comprise clocking logic which provides a delay of one chip time Tcto the half symbol code from the Q-lookup table.

Once a set of 32 unique symbol codes are selected, the contents of theI-lookup table and Q-lookup table can be generated by dividing eachsymbol code into even and odd chips, and using the even chips for thehalf symbol codes in the Q-lookup table and the odd chips for the halfsymbol codes in the I-lookup table. Other techniques for generating evenand odd chip sequences suitable for signals q(t) and i(t) fall withinthe spirit and scope of the invention.

FIG. 7 is a block diagram of a spread spectrum receiver.

The transmitted signal s(t) 313 may undergo attenuation, addition ofnoise, multipath superposition, and other known and unknown effects ofthe transmission channel 107. Accordingly, the received signal s*(t) 401may differ from the transmitted signal s(t) 313 in known and unknownways.

Received signal s*(t) may be despread using multiple correlators keyedto I and Q chip streams. Because CPM spread spectrum signals may bethought of as the superposition of time staggered signals created from Iand Q chip streams (each at half the chip rate), a receiver according toone embodiment of the present invention uses two correlators, oneprogrammed with the I-chip-sequence and one programmed with theQ-chip-sequence and both operating at half the chip rate, to decode thereceived signal, and then combines the outputs of the two correlators.

In the receiver of FIG. 7, the received signal s*(t) 401 is coupled to aCPM correlator 402 for recognizing a chip sequence in the receivedsignal s*(t) 401. The CPM correlator 402 comprises a power divider 403for generating duplicate signals, an i*(t) signal 404 with a 0 degreephase delay, and a q*(t) signal 405 with a 90 degree phase shift.

The i*(t) signal 404 is coupled to a delay 406, which delays the i*(t)signal 404 by one chip time Tc to allow simultaneous generation ofcorrelation pulses by the I correlator 407 and the Q correlator 409.Thus, the delayed i*(t) signal is coupled to an I correlator 407, andthe q*(t) signal 405 is coupled directly to a Q correlator 409.

The I correlator 407 operates at a chip rate of Rc/2, where Rc is thechip rate (i.e., 1/Tc) of the received signal s*(t) 401. The Icorrelator 407 may comprise one of several types of correlators, e.g., asurface-acoustical-wave (SAW) correlator, a tapped-delay-line (TDL)correlator, or a serial correlator. Examples of suitable correlators maybe found in U.S. Pat. No. 5,016,255 entitled “Asymmetric Spread SpectrumCorrelator” or in U.S. Pat. No. 5,022,047 entitled “Spread SpectrumCorrelator”, both of which are issued in the name of inventors Robert C.Dixon and Jeffrey S. Vanderpool and hereby incorporated by reference asif fully set forth herein. The I correlator 407 produces an output Icorrelation signal 408 indicating a degree of match between the delayedi*(t) signal and a predetermined I-chip-sequence.

The Q correlator 409 similarly operates at a chip rate of Rc/2, and maysimilarly comprise any of a number of suitable correlators such as thosedescribed in the patents referenced in the preceding paragraph. The Qcorrelator 409 produces an output Q correlation signal 410 indicating adegree of match between the q*(t) signal and a predeterminedQ-chip-sequence.

The I correlation signal 408 and the Q correlation signal 410 arecoupled to a summer 411, which combines its inputs and produces aunified correlation signal 412. Because the i*(t) signal is delayed bydelay 406, the I correlation signal 408 and Q correlation signal 410occur simultaneously. The unified correlation signal 412 is used todetermine a data stream d(t) from which the chip sequence c(t) wasgenerated.

The I correlator 407 and the Q correlator 409 thus jointly identify thechip sequence in the received signal s*(t) 401. The I correlator 407 isconfigured to recognize the odd chips of the chip sequence, while the Qcorrelator 409 is configured to recognize the even chips of the chipsequence. When the entire correlation sequence appears in the receivedsignal s*(t), the sum of the I correlation signal 408 and the Qcorrelation signal 410 is at a maximum, and may be compared against apredetermined threshold to allow recognition of the chip sequence. Aunified correlation signal 412 is produced when a chip sequence isrecognized.

Alternatively, instead of comparing the unified correlation signal 412to a predetermined threshold, a system may be configured so as to have aplurality (e.g., 32) of CPM correlators 402 operating in parallel, eachtuned to recognize a different code sequence. The outputs of all 32 CPMcorrelators may be summed and, when the sum is at a predeterminedmaximum level, the CPM correlator 402 with the highest magnitude outputmay be chosen by a best-of-M detector or similar means as indicative ofthe data stream d(t). For example, in a CSK system, each of 32 CPMcorrelators may attempt in parallel to recognize a code sequence, andthe one with the highest magnitude correlation signal may be assumed toindicate the received chip stream. The recognized chip stream maycorrespond to a data symbol from which a portion of the data stream d(t)may be recovered.

In a preferred embodiment, the CPM correlator 402 may be used inconjunction with techniques described in U.S. Pat. Nos. 5,016,255 or5,022,047, both of which are assigned to the assignee of the presentinvention and hereby incorporated by reference. In those techniques,each data bit or data symbol of the data stream d(t) may be encoded bymodulation with the entire length of a pseudo random chip sequencegenerated from a chip sequence code. For example, if a chip sequencecode identifies a pseudo random chip sequence that repeats after 32chips, each data bit of the data stream d(t) may be modulated with all32 of those chips.

However, there is no requirement that the CPM correlator 402 be usedwith those particular techniques. For example, the CPM correlator may beused with other spread spectrum techniques to recognize a correlationsignal that is used to synchronize the transmitter 101 and the receiver108. Also, the CPM correlator 402 may be used in conjunction with aself-synchronizing or auto-synchronizing spread spectrum technique suchas described elsewhere herein in more detail.

The I and Q chip sequences are preferably of equal length; thus, eachCSK symbol code is preferably an even number of chips in length so as toavoid a 90-degree phase uncertainty between symbol codes whendespreading is attempted.

FIG. 8 is a block diagram of a coherent spread spectrum receiver.

The received signal s*(t) 401 in the receiver of FIG. 8 is coupled to aCPM correlator 502 for recognizing a chip sequence in the receivedsignal s*(t) 401. The CPM correlator 502 comprises a power divider 503,which produces duplicate signals 504 and 505, each with a 0 degree phasedelay. Such power dividers are known in the art and are generallypreferred for the CPM correlator 502 over the power divider 403 shown inFIG. 7. While a phase delay of 90 degrees between i*(t) and q*(t) wasimposed by use of the power divider 403 in FIG. 7, a 90-degree phasedelay in the FIG. 8 embodiment is produced by separately multiplying thesignals 504 and 505 with cosine and sine signals, respectively.

The signal 504 is multiplied with a cos w₀t signal by I multiplier 530and filtered by a I low pass filter 506 to provide an i*(t) signal. Thesignal 505 is multiplied by a sin ω₀t signal by Q multiplier 531 andfiltered by a Q low pass filter 512 to provide a q*(t) signal.

The outputs of the I low pass filter 506 and the Q low pass filter 512generally appear for MSK as half sinusoidal waveforms corresponding tothose generated in the transmitter from P(t) generators 305, 306.

The i*(t) signal output from I low pass filter 506 is coupled to an Icorrelator 507. The I correlator 507 comprises a register 508 having asequence of chips 509. The register 508 may be an analog shift register,a tapped delay line having a plurality of taps, or any other suitablestorage means. The odd chips are coupled by a plurality of multipliersto an I summer 510, which combines its inputs and produces an output Icorrelation signal 511.

An example of the path of the I correlator 507 is shown in FIG. 23. Asdescribed with respect to FIG. 8, the filtered i*(t) signal is coupledto a register 508. The register 508 comprises a series of chips 509along which the filtered i*(t) signal propagates. The register 508 ismatched to a particular code sequence. Thus, in the example of FIG. 23,the sequence of odd chips which will result in a match isC_(odd)(t)=11001000. At time t=16Tc, the first chip C₁ is compared withthe first chip in the sequence of C_(odd)(t), and a “1” is generated ifthe chips are equal. Each of the other odd chips in the register 508 islikewise compared against the programmed sequence. A comparison betweenany two chips may be carried out using a multiplier or an exclusive-ORgate. The comparison values are provided to a summer 510 which generatesa maximum pulse when the chip sequence for which the correlator 507 hasbeen programmed matches the received chip sequence. In FIG. 23, thebranches having a “−1” correspond to chips for which a “0” in thereceived chip sequence will generate a match, while the other branchescorrespond to chips for which a “1” in the received chip sequence willgenerate a match.

Returning to FIG. 8, the q*(t) signal output from the Q low pass filter512 is coupled to a Q correlator 513. The Q correlator 513 similarlycomprises a register 514 having a sequence of chips 515. As with the oddchips in the I correlator 507, the even chips are coupled to a Q summer516, which combines its inputs and produces an output Q correlationsignal 517.

The I correlation signal 511 and the Q correlation signal 517 arecoupled to a summer 518, which combines its inputs and produces aunified correlation signal 519. Because the I correlation signal 511 isderived from the odd chips while the Q correlation signal 517 is derivedfrom the even chips (which precede the odd chips by one chip time Tc),the correlation signals 511, 517 occur simultaneously, and there is noneed for a separate delay element such as delay 406 shown in FIG. 7. Theunified correlation signal 519 is used to determine a data stream d(t)from which the chip sequence c(t) was generated in a manner similar tothat explained above with reference to FIG. 7.

The FIG. 8 receiver operates best with a coherent carrier reference ω₀and assumes such is available. Methods are known in the art forobtaining a coherent carrier reference, such as the use of phaseestimating circuitry. Where very rapid acquisition times are necessary,such as in certain high-speed time division multiple access (TDMA)systems using CPM spread spectrum techniques, other embodiments (such asthe non-coherent receiver embodiments described herein) may generally bepreferred.

In a non-coherent CPM system, the receiver 108 of FIG. 1 may not haveavailable an exact copy of the carrier signal at frequency ω₀ used bythe transmitter 101. Rather, the receiver 108 generates a local carriersignal having a frequency ω₁, which in practice may differ in frequencyand phase from the transmitter's carrier signal:

cos ω₁t=cos(ω₀+Δω)t+θ  (603)

where Δω=frequency difference and θ=phase difference.

FIG. 10 is a block diagram of a non-coherent spread spectrum receiverfor receiving and despreading a CPM spread spectrum signal without theneed for a locally generated coherent reference signal ω₀. The receiverof FIG. 10 can be used to process a received CPM signal by splitting thereceived spread spectrum signal into separable real and imaginary parts,splitting the real and imaginary parts into I and Q portions, mixing thereal I and Q portions and the imaginary I and Q portions with anon-coherent reference signal having a frequency near that expected ofthe received signal to obtain real I and Q streams and imaginary I and Qstreams, filtering the multiplied signals, correlating separately the Iand Q streams for each of the real and imaginary parts to obtain a realI and Q correlation pulse and an imaginary I and Q correlation pulse,combining the I and Q correlation pulses separately for the real andimaginary parts to provide a combined real and a combined imaginarycorrelation signal, squaring the combined real and imaginary correlationsignals to generate a squared real and a squared imaginary correlationpulse, and combining the squared real and imaginary correlation signalsinto a unified correlation signal.

The operation of the receiver of FIG. 10 may be explained graphicallywith reference to FIG. 9, which is a scatter diagram comparing real andimaginary values as transmitted and as received in a non-coherentreceiver. For simplicity, the explanation below assumes the transmissionchannel to be distortionless and have unlimited bandwidth. Thetransmitter's coordinate system 601 is represented by an x-axis andy-axis, with the x-axis representing values of i(t) and the y-axisrepresenting values of q(t). A set of four points 610 through 613represents transmitted sampled value pairs for <i(t_(n)),q(t_(n))>. Thepairs 610 through 613 represent coordinates <1,0>, <0,1>, <−1, 0>, and<0,−1>, respectively.

A receiver's coordinate system 604 is represented by an x*-axis and ay*-axis shown as dashed lines in FIG. 9. The receiver's coordinatesystem 604 is assumed to differ from the transmitter's coordinate system601 due to frequency and phase differences. The receiver's coordinatesystem 604 rotates with respect to the transmitter's coordinate system601 at a rate proportional to Δω, the frequency difference (“beatfrequency”) between the transmitter and receiver reference signals. Forsufficiently small Δω (such as may be expected for the time period ofinterest over which correlation for a data symbol will occur—e.g., 32chip periods), the receiver's coordinate system 604 approximately equalsthe transmitter's coordinate system 601, except for a phase difference θwhich remains relatively constant for short periods of time. In order tomaintain such a condition, the beat frequency Δω preferably should beless than about ¼ the symbol rate. For example, with a symbol rate of156.25k symbols/second (5 Mchips/second), the beat frequency Δω shouldbe less than about 39 kHz for optimal operation.

Because the receiver's coordinate system 604 at a given instant appearsrotationally shifted with respect to the transmitter's coordinate system601, the <i*(t_(n)),q*(t_(n))> sampled pair recognized by the receiver108 will be a point on the circle 607 corresponding to an<i(t_(n)),q(t_(n))> sampled pair in the transmitter's coordinate system601 but shifted around circle 607 by an amount dependent on the phasedifference θ. Accordingly, the perceived real value or i*(t) will differfrom the transmitted i(t) value by an amount dependent upon cos θ due tothe rotational difference between the coordinate systems 601 and 604,while the perceived imaginary value or q*(t) will also differ from thetransmitted q(t) value by an amount dependent upon sin θ for the samereason. Thus, if the transmitted <i(n), q(n)> sampled values are <1, 0>and the phase offset θ is +30°, the received <i*(t_(n)), q*(t_(n))>sampled values are <cos +30°, sin +30°> or <0.866, 0.5>. Likewise, ifthe phase offset θ is +90° for the same transmitted values, the received<i*(t_(n)), q*(t_(n))> sampled values are <0, 1>.

From the above explanation, it is apparent that a correlator attemptingto correlate for I and Q portions would be faced with a diminishingi*(t) value as θ varies from 0 to 90 degrees, yet at the same time anincreasing q*(t) value. As θ grows, eventually the difference between<i(t), q(t)> and <i*(t), q*(t)> becomes so large that accuratecorrelation is cumbersome. Because of the phase difference θ, it isgenerally not possible to know in advance which quadrant of FIG. 9 thereceived signal s*(t) will be in relative to the transmitter'scoordinate system 601. However, the present invention in one aspectaddresses this problem by utilizing both real and imaginary parts of Iand Q portions in order to despread the received s*(t) signal.

It may be noted that as the real portion of i*(t) decreases as θ variesfrom 0 to 90 degrees, the imaginary portion of i*(t) increases.Similarly, as the real portion of i*(t) increases (in magnitude) as θvaries from 90 to 180 degrees, the imaginary portion of i*(t) decreases.A similar phenomenon occurs with the real and imaginary portions ofq*(t). The receiver of FIG. 10 takes advantage of the complementaryaspects of the real and imaginary portions of the received i*(t) andq*(t) signal portions, and accordingly analyzes both the real andimaginary parts of the I and Q signals in order to make an effectivecorrelation.

In the FIG. 10 embodiment, the received signal s*(t) 401 is coupled to anon-coherent CPM correlator 702 for recognizing a correlation sequencein the received signal s*(t) 401. The non-coherent CPM correlator 702comprises a power divider 703, which produces duplicate signals Real*(t)704 having a 0-degree phase delay and Imag*(t) 705 having a 90-degreephase delay. Real*(t) 704 and Imag*(t) 705 may be viewed as the real andimaginary parts of the received signal s*(t) 401.

The Real*(t) signal 704 is coupled to a CPM correlator 715 similar toCPM correlator 502 of FIG. 8, with the exception that the localreference signal is different, as described below. The CPM correlator715 produces a real correlation signal 706. The Imag*(t) signal iscoupled to a second CPM correlator 715 which produces an imaginarycorrelation signal 707. The real correlation signal 706 is coupled to asquaring device 708, which computes the square of its input. Theimaginary correlation signal 707 is likewise coupled to a squaringdevice 709, which computes the square of its input. The outputs of thesquaring devices 708 and 709 are coupled to a summer 710, which combinesits inputs to produce a unified correlation signal 711 which is the sumof the squares of the real correlation signal 706 and the imaginarycorrelation signal 707. The unified correlation signal 711 is coupled toa square root device 712 which takes the square root of its input, andgenerates a final correlation signal 713 comprising correlation pulses714. The time between correlation pulses 714 may be one symbol code timeperiod Ts if CSK is employed.

A primary difference between the CPM correlators 715 shown in FIG. 10and the CPM correlator 502 of FIG. 8 is that the CPM correlators 715 ofFIG. 10 utilize non-coherent reference signals cos ω₁t=cos(ω₀+Δω)t+θ andsin ω₁t=sin(ω₀+Δω)t+θ for the I and Q portions, respectively, ratherthan cos ω₀t and sin ω₀t as generated in the coherent receiver of FIG.8. The reference signals cos ω₁t and sin ω₁t may be generated from thesame oscillator coupled to a power divider to keep the phase offset θthe same for both cos ω₁t and sin ω₁t. The use of non-coherent referencesignals causes the correlation signal generated by each CPM correlator715 to have a magnitude dependent in part upon the phase difference θ.

The effect of using non-coherent reference signals on the ability toachieve correlation may be explained first with reference to the Iportion of the Real*(t) signal 704. The Real*(t) signal 704 may berepresented as:

Real*(t)=Re{A u(t)exp(−jω₀t)}

where, as mentioned previously, u(t)=i(t)+jq(t), which is the complexenvelope of s(t), and Re {} denotes the real portion of a complex value.The Real*(t) signal 704 is multiplied by multiplier 720 with a locallygenerated reference signal cos ω₁t=cos(ω₀+Δω)t+θ, so that the output ofmultiplier 720 is:

Re {A u(t)exp(−jω₀t)}cos ω₁(t)

The output of the multiplier 720 is coupled to a low pass filter 721which retains the baseband portion of the signal coupled to its input.Assuming that the non-coherent reference signal cos ω₁t differs from thetransmitter reference frequency ω₀ by only a phase difference (i.e.,that the frequency change is negligible over the time period ofinterest), then the receiver reference signal may be expressed as:

cos ω₁t=cos(ω₀t+θ)

The output y(t) of the low pass filter 721 may therefore be expressedas:

y(t) = LPF [ Re {A u(t) exp(−jω₀t)} cos ω₁(t) ] = LPF [ Re {A u(t)exp[j(−ω₀t + ω₁t)]} ] = (A/2) i(t) cos(ω₀ + ω₁t)t + (A/2) q(t) sin(ω₀t +ω₁t) = (A/2) i(t) cos(−θ) + (A/2) q(t) sin (−θ) = (A/2) i(t) cos θ −(A/2) q(t) sin θ (790)

where “LPF” denotes operation of the low pass filter 721.

By similar deduction the output z(t) of the low pass filter 731 of the Qportion of the Real*(t) signal is as follows:

z(t) = (A/2) i(t) sin(−θ) + (A/2) q(t) cos(−θ) = (−A/2) i(t) sin θ +(A/2) q(t) cos θ (791)

Due to the 90-degree phase shift in signal 705, the output of low passfilter 741 of the I portion of the Imag*(t) signal is equal to z(t) asderived above, while the output of low pass filter 743 of the Q portionof the Imag*(t) signal is equal to the inverse of y(t) as derived above.

In operation, each of the four correlators 722 through 725 maycontribute to correlation of the received CPM signal s*(t). Operation ofthe non-coherent CPM correlator 702 may be shown through severalexamples. As a first example, assume that the phase offset θ=0°;therefore, the outputs y(t) and z(t) for low pass filters 721 and 731,respectively, reduce to the following:

y(t)=(A/2) i(t)

and

z(t)=(A/2) q(t)

Selecting an amplification factor A=2, the filter outputs of filters 721and 731 then become y(t)=i(t) and z(t)=q(t). Assuming, for convenience,a code sequence length of 16 chips, then after 16 chip times (i.e.,16Tc) the entire sequence is contained within the correlation registers726, 727, 728, and 729 in each CPM correlator 715. An illustrative chipstream c(t)=1111010110010000 may be broken into sub-sequencesC_(odd)(t)=11001000 and C_(even)(t)=11110100. It will further be assumedfor sake of explanation that the waveform generator P(t) of thetransmitter generates a return-to-zero (RZ) rectangular waveform havinga duration of two chip periods, so that the transmitted i(t) and q(t)signals may be depicted as shown in FIG. 11A and FIG. 11B, respectively.Operation of the FIG. 10 correlator using CPM baseband signals insteadof RZ signals can be understood by observing that at time t=16Tc, thepeak values of the sinusoidal waveforms appear in the correlationregisters 726, 727, 728 and 729, and correspond to the pulse height ofthe RZ waveform.

At the receiving end, the contents of the correlation registers 726 and727 may be represented as shown in FIGS. 11C and 11D, respectively. Itcan be seen that the waveform of FIG. 11C as reading from right to leftis the same as that of FIG. 11A as reading from left to right.Similarly, the waveforms of FIGS. 11B and 11D bear the samerelationship. An output for each of the four correlators 722, 723, 724and 725 may be obtained by pointwise multiplication of the chip valueswith the chip weighting factors 716 for each chip, and summation of thechip products by summers 717 to produce a correlation signal. The chipweighting factors 716 for correlator 725 are opposite in sign to thevalues for correlator 723. The chip weighting factors 716 forcorrelators 722 and 724 are the same sign.

Continuing with the present example in which θ=0°, the output at timet=16Tc for each of correlators 722 and 723, corresponding respectivelyto the I portion (“ReI”) and the Q portion (“ReQ”) of the Real*(t)signal, is eight, while the output for each of correlators 724 and 725,corresponding respectively to the I portion (“ImI”) and the Q portion(“ImQ”) of the Imag*(t) signal, is 0. The final correlation signal 713at the instant 16Tc is: $\begin{matrix}{\begin{matrix}{{Corr}\left( {t = {16{Tc}}} \right)}\end{matrix} = \quad \left\{ {\left( {{{Re}I} + {{Re}Q}} \right)^{2} + \left( {{{Im}I} + {{Im}Q}} \right)^{2}} \right\}^{1/2}} \\{= \quad {\left\{ \left( {8 + 8} \right)^{2} \right\}^{1/2} = 16}}\end{matrix}$

The value of 16 is a maximum value indicating correlation for theparticular chip sequence. If multiple codes are to be recognized, aplurality of non-coherent CPM correlators 702 may operate in parallel,each programmed to recognize a different code. The chip sequencecorresponding to the highest correlation signal may be selected as thereceived chip sequence.

Assuming as a second example that θ=30°, the contents of correlationregisters 726 and 727 appear as shown in FIGS. 11E and 11F,respectively. Selecting the amplification factor A=2, the outputs y(t)and z(t) of low pass filters 721 and 731, respectively, may berepresented as: $\begin{matrix}{{y(t)} = {{\left( {A/2} \right){i(t)}{\cos \left( {30{^\circ}} \right)}} - {\left( {A/2} \right){q(t)}{\sin \left( {30{^\circ}} \right)}}}} \\{= {{{i(t)}(0.866)} - {{q(t)}(0.5)}}}\end{matrix}$ and $\begin{matrix}{{z(t)} = {{\left( {{- A}/2} \right){i(t)}{\cos \left( {30{^\circ}} \right)}} + {\left( {A/2} \right){q(t)}{\sin \left( {30{^\circ}} \right)}}}} \\{= {{{- {i(t)}}(0.5)} + {{q(t)}(0.866)}}}\end{matrix}$

Pointwise vector multiplication of each of the chip valves in thecorrelation registers 726 through 729 with corresponding chip weights716 yields the following outputs from summers 717:

ReI = (1)(0.866) + (1)(0.866) + (−1)(−0.866) + (−1)(−0.866) . . . =(8)(0.866) = 6.928 ReQ = (1)(0.866) + (1)(0.866) + (1)(0.866) . . . =(8)(0.866) = 6.928 ImI = (1)(−0.5) + (1)(−0.5) + (−1)(0.5) + (−1)(0.5) .. . = −(8)(0.5) = −4.0 ImQ = (1)(−0.5) + (1)(−0.5) + (1)(−0.5) +(1)(−0.5) . . . = −(8)(0.5) = −4.0

A final correlation signal 713 therefore is generated:

Corr(t=16Tc)={(6.928+6.928)²+(−4+−4)²}^(½)=16

Thus, for a phase offset of θ=30°, the value of the final correlationsignal 713 at t=16Tc remains at the maximum level of 16.

As another example, a phase offset θ=45° is assumed. The outputs y(t)and z(t) of low pass filters 721 and 731, respectively, become:

y(t)=i(t)(0.707)−q(t)(0.707)

and

z(t)=−i(t)(0.707)+q(t)(0.707)

Solving for the intermediate values ReI, ReQ, ImI, and ImQ yields:

ReI = (1)(0.707) + (1)(0.707) . . . = (8)(0.707) = 5.657 ReQ =(1)(0.707) + (1)(0.707) . . . = (8)(0.707) = 5.657 ImI = (1)(−0.707) +(1)(−0.707) . . . = −(8)(0.707) = −5.657 ImQ = (1)(−0.707) + (1)(−0.707). . . = −(8)(0.707) = −5.657

A final correlator signal 713 is generated:

Corr(t=16Tc)={(2×5.657)²+(2×−5.657)²}^(½)=16

Again, maximum correlation of 16 is realized even though the phaseoffset θ is not equal to 0.

A table can be constructed of (ReI+ReQ), (ImI+ImQ) values andcorrelation values versus phase offset θ for the correlator of FIG. 10:

θ R_(i) + R_(q) I_(i) + I_(q) Corr=  0° 16 0.0 16.0  30 13.856 −8.0 16.0 45 11.314 −11.314 16.0  60 8.0 −13.856 16.0  90 0.0 −16.0 16.0 120 −8.0−13.856 16.0 135 −11.314 −11.314 16.0 150 −13.856 −8.0 16.0 180 −16.00.0 16.0 210 −13.856 8.0 16.0 225 −11.314 11.314 16.0 240 −8.0 13.85616.0 270 0.0 16.0 16.0 300 8.0 13.856 16.0 315 11.314 11.314 16.0 33013.856 8.0 16.0

As the phase offset a increases beyond 45°, a higher percentage of thecorrelation value begins to come from the Imag*(t) signal path 705rather than the Real*(t) signal path 704 of the non-coherent CPMcorrelator 702. At a phase offset of θ=90°, for example, all correlationis coming from the Imag*(t) signal path 705 and none from the Real*(t)signal path 704. The output 706 of the real CPM correlator 715 andoutput 707 of the imaginary CPM correlator 715 progress sinusoidally asa function of the phase offset θ and can be expressed as:

Real*(t) correlation=16 cos θ

Imag*(t) correlation=−16 sin θ

Corr={(16 cos θ)²+(−16 sin θ)²}^(½)=16

Thus, maximum correlation of 16 will be achieved regardless of the phaseoffset θ. The use of multiple correlators as configured in the mannershown in FIG. 10 allows successful correlation regardless of whichquadrant of FIG. 9 the receiver operates with respect to thetransmitter.

It should be noted that at chip times other than multiples of 16Tc (forthe example of chip sequence of 16 chips), the correlation output willbe a function of the cross correlation value between the i(t_(n)) andq(t_(n)) subcodes. The non-coherent CPM correlator of FIG. 10 shouldperform no worse as far as cross-correlation than a bi-phase correlatorwith the same code. In other words, if a given code produces a maximumtime sidelobe value of 4/16 through bi-phase correlation, then the worsttime sidelobe to be expected from the FIG. 10 correlator should also be4/16.

FIG. 12 is a block diagram of another embodiment of a non-coherentspread spectrum correlator using separable real and imaginary parts ofthe received spread spectrum signal. The FIG. 12 correlator uses onlytwo shift registers instead of four shift registers and uses only asingle power divider having no imposed phase delay for operating on thereceived signal s*(t) as opposed to three power dividers in thenon-coherent correlator illustrated in FIG. 10. The use of a powerdivider having no imposed phase delay on the received signal s*(t) is anadvantage because power dividers which impose a phase delay on thereceived signal typically operate optimally over only a relativelynarrow bandwidth, while the received signal may cover a relatively widebandwidth.

In FIG. 12, the received signal s*(t) 401 is coupled to a two-registernon-coherent CPM correlator 802 for recognizing a chip sequence in thereceived signal s*(t). The two-register non-coherent CPM correlator 802comprises a first power divider 803, which produces duplicate signals804 and 805, each with a 0-degree phase delay. A local oscillator 806produces a local carrier signal cos ω₁t 807, which is coupled to asecond power divider 808. The second power divider 808 producesduplicate signals, one signal 809 with a 0-degree phase delay, andanother signal 810 with a 90-degree phase delay. The use of the secondpower divider 808 to generate signals cos ω₁ and sin ω₁ from the samelocal oscillator 806 maintains the phase offset θ between ω₁ and ω₀ forboth cos ω₁ and sin ω₁.

The signals 804 and 809 are coupled to a first multiplier 811, whichcombines its inputs and produces a first product signal 812. The firstproduct signal 812 is coupled to a first low pass filter 813, whichproduces a first filtered signal 814 which retains its basebandfrequency components. The first filtered signal 814 is coupled to afirst even-odd correlator 815.

The signals 805 and 810 are similarly coupled to a second multiplier816, which combines its inputs and produces a second product signal 817.The second product signal 817 is similarly coupled to a second low passfilter 818, which produces a second filtered signal 819 which retainsits baseband frequency components. The second filtered signal 819 issimilarly coupled to a second even-odd correlator 820.

In the two-register non-coherent CPM correlator 802 depicted in FIG. 12,the Q portion of the Real*(t) signal is the same as the I portion of theImag*(t) signal, and the Q portion of the Imag*(t) signal is 180-degreesout of phase (i.e., the inverse) of the I portion of the Real*(t)signal. The Q portion of the Real*(t) signal and the I portion of theImag*(t) signal are stored in and read from the same register 821.Similarly, the Q portion of the Imag*(t) signal and the I portion of theReal*(t) signal are stored in and read from the same register 827. Thetwo-register non-coherent CPM correlator 802 of FIG. 12 operates in aconceptually similar manner to the non-coherent CPM correlator 702 ofFIG. 10.

The first even-odd correlator 815 simultaneously recognizes the reali*(t) components and the imaginary q*(t) components, and comprises aregister 821 capable of holding a sequence of chips 822. The odd chipsare coupled to a real I summer 823, which combines its inputs andproduces a real I correlation signal 824. The even chips are coupled toan imaginary Q summer 825, which combines its inputs and produces animaginary Q correlation signal 826.

The second even-odd correlator 820 simultaneously recognizes theimaginary i*(t) components and the real q*(t) components, and comprisesa register 827 capable of holding a sequence of chips 828. The odd chipsare coupled to an imaginary I summer 829, which combines its inputs andproduces an imaginary I correlation signal 830. The even chips arecoupled to a real Q summer 831, which combines its inputs and produces areal Q correlation signal 832.

The real I correlation signal 824 and the real Q correlation signal 832are coupled to a real summer 833, which combines its inputs and producea real correlation signal 834. Similarly, the imaginary Q correlationsignal 826 and the imaginary I correlation signal 830 are coupled to animaginary summer 835, which combines its inputs and produces animaginary correlation signal 836.

The real correlation signal 834 is coupled to a squaring device 837,which computes the square of its input. The imaginary correlation signal836 is coupled to a squaring device 838, which computes the square ofits input. The two squared values are coupled to a summer 839, whichcombines its inputs and produces a unified correlation signal 840representing the sum of the squares of the real correlation signal 834and the imaginary correlation signal 836. The unified correlation signal840 is coupled to a square root device 841 which takes the square rootof its input and generates a final correlation signal 842. The squaringdevices 837 and 838, the summer 839, and the square root device 841collectively compute the root of the sum of the squares of the real andimaginary signals. A Robertson device such as depicted in FIG. 22 anddescribed elsewhere herein may be used to estimate the root of the sumof the squares. The time between separate correlation pulses 843 may beone symbol code time period Ts if CSK is used.

It should be noted that in the non-coherent CPM correlator 702 of FIG.10 and the two-register non-coherent CPM correlator 802 of FIG. 12, theprocess of squaring destroys polarity information.

FIG. 13A is a block diagram of a spread spectrum receiver using serialcorrelation.

The received signal s*(t) 401 is coupled to a coherent serial CPMcorrelator 902 for recognizing a correlation sequence in the receivedsignal s*(t) 401.

The coherent serial CPM correlator 902 of FIG. 13A comprises a powerdivider 903, which produces duplicate signals 904 and 905 with a0-degree phase delay. The signal 904 is coupled to an I multiplier 906.The other input of the I multiplier 906 is coupled to a locallygenerated signal i(t) cos ω₀t that is, the carrier signal combined withthe I chip sequence of the correlation sequence. The signal 905 iscoupled to a Q multiplier 911, which is coupled to a locally generatedsignal q(t) sin ω₀t, that is, the carrier signal combined with the Qchip sequence of the correlation sequence. The coherent serial CPMcorrelator of FIG. 13A uses a coherent reference signal having afrequency ω₀.

The i(t) signal, which is the waveform representing the I chip sequence,and the q(t) signal, which is the waveform representing the Q chipsequence, each comprise tri-valued return to zero (RZ) waveforms, thatis, they are +1 to indicate a logical “1”, −1 to indicate a logical “0”,and 0 to indicate no value, as shown in FIG. 13B. The i(t) signal andthe q(t) signal are offset by one chip time from each other in the sensethat the i(t) signal has a value of +1 or −1 at each odd chip time butis 0 during the even chip times, and the q(t) signal has a value of +1or −1 at each even chip time but is 0 during the odd chip is times.

The I multiplier 906 combines its inputs and produces an I productsignal 907. The I product signal 907 is filtered by a low pass filter(not shown) and is coupled to an I integrator 908, which integrates itsinput and dumps the sum under control of a control input 909. The Iintegrator 908 produces an I correlation signal 910.

The Q multiplier 911 combines its inputs and produces a Q product signal912. The Q product signal 912 is filtered by a low pass filter (notshown) and coupled to a Q integrator 913, which integrates its input anddumps the sum under control of a control input 914. The Q integrator 913produces a Q correlation signal 915. Because the i(t) signal and theq(t) signals are tri-valued return to zero waveforms, only one of theintegrators 908, 913 changes value at a time.

The I correlation signal 910 and the Q correlation signal 915 arecoupled to a summer 916, which combines its inputs and produces aunified correlation signal 917. The unified correlation signal 917increases progressively in a stepwise fashion and reaches a maximum whenfull correlation is achieved. If CSK is used, then the largest of theunified correlation signals 917 for a plurality of parallel coherentserial CPM correlators 902 over a given symbol code time Ts may be usedto identify the received symbol code. The I and Q integrators 908, 913hold their values until instructed to dump.

To properly control the integrate and dump operation, synchronizationinformation is necessary. To accomplish this, a parallel correlator mayoperate in conjunction with one or more serial correlators to providethe necessary timing information. In such an embodiment, a transmittermay first transmit data (e.g., a preamble) which is received by theparallel correlator. The parallel correlator generates a correlationpulse when the received data is recognized, which correlation pulse isused to control the timing of the serial correlator or correlators.

FIG. 14 is a block diagram of a non-coherent spread spectrum receiverusing serial correlation for separable real and imaginary parts of thereceived spread spectrum signal.

Conceptually, the non-coherent serial CPM correlator of FIG. 14 operatesin a similar fashion as the non-coherent CPM correlator 702 of FIG. 10.The received signal s*(t) 401 is coupled to a non-coherent serial CPMcorrelator 1002 for recognizing a chip sequence in the received signals*(t) 401. The non-coherent serial CPM correlator 1002 comprises a powerdivider 1003, which produces duplicate signals, Real*(t) 1004 having a0-degree phase delay, and Imag*(t) 1005 having a 90-degree phase delay.Real*(t) 1004 and Imag*(t) 1005 are the real and imaginary parts of thereceived signal s*(t) 401.

The Real*(t) signal 1004 is coupled to a serial CPM correlator 1020which produces a real correlation signal 1006. The Imag*(t) signal 1005is similarly coupled to a second serial CPM correlator 1020 whichproduces an imaginary correlation signal 1007.

Each serial CPM correlator 1020 comprises a power divider (not shown)which receives an input signal and splits it into duplicate signals 1021and 1022 with a 0-degree phase delay. The signal 1021 is coupled to afirst I multiplier 1023. The other input of the first I multiplier 1023is coupled to a locally generated non-coherent reference signal cosω₁t=cos(ω₀+Δω)t+θ as described earlier with reference to FIG. 10. Theoutput of the first I multiplier 1023 is coupled to an I low pass filter1027, the output of which is coupled to a second I multiplier 1029. Theother input of the second I multiplier 1029 is coupled to an i(t) signal1031, which is the waveform representing the I chip sequence (see FIGS.13A and 13B).

The signal 1022 is coupled to a first Q multiplier 1024. The other inputof the first Q multiplier 1024 is coupled to a locally generatednon-coherent reference signal sin ω₁t=sin(ω₀+Δω)t+θ as described earlierwith reference to FIG. 10. The output of the first Q multiplier 1024 iscoupled to a Q low pass filter 1028, the output of which is coupled to asecond Q multiplier 1030. The other input of the second Q multiplier1030 is coupled to a q(t) signal 1032, which is the waveformrepresenting the Q chip sequence (see FIGS. 13A and 13B).

The output of the second I multiplier 1029 is coupled to an I integrator1033, which integrates its input and dumps the sum under control of acontrol input 1035. The I integrator 1033 produces an I correlationsignal 1037.

The output of the second Q multiplier 1030 is coupled to a Q integrator1034, which integrates its input and dumps the sum under control of acontrol input 1036. The Q integrator 1034 produces a Q correlationsignal 1038.

The i(t) signal, which is the waveform representing the I chip sequence,and the q(t) signal, which is the waveform representing the Q chipsequence, each comprise tri-valued return to zero (RZ) waveforms, thatis, they are +1 to indicate a logical “1”, −1 to indicate a logical “0”,and 0 to indicate no value, as shown in FIG. 13B. The i(t) signal andthe q(t) signal are offset by one chip time from each other in the sensethat the i(t) signal has a value of +1 or −1 at each odd chip time butis 0 during the even chip times, and the q(t) signal has a value of +1or −1 at each even chip time but is 0 during the odd chip times. Becausethe i(t) signal and the q(t) signals are tri-valued return to zerowaveforms, only one of the integrators 1035, 1036 changes value at atime. The I and Q integrators 1035, 1036 hold their values untilinstructed to dump.

As noted with respect to FIG. 13A, synchronization information necessaryfor controlling the integrate and dump operation of the I and Qintegrators 1035, 1036 may be obtained from a parallel correlatorreceiving timing information from a transmitted preamble in order togenerate a correlation pulse. The correlation pulse may be used tocontrol the timing of the serial correlator or correlators. Othersuitable methods of control are also possible.

The I correlation signal 1037 and the Q correlation signal 1038 arecoupled to a summer 1039, which combines its inputs and produces aunified correlation signal 1006. The unified correlation signal 1006increases progressively in a stepwise fashion and reaches a maximum whenfull correlation is achieved. As noted, the CPM correlator 1020receiving the Real*(t) signal 1004 produces a real correlation signal1006, and the second CPM correlator 1020 receiving the Imag*(t) signal1005 produces an imaginary correlation signal 1007.

The real correlation signal 1006 is coupled to a squaring device 1008,which computes the square of its input. The imaginary correlation signal1007 is coupled to a squaring device 1009, which computes the square ofits input. The two squared values are coupled to a summer 1010, whichcombines its inputs and produces a unified correlation signal 1011representing the sum of the squares of the real correlation signal 1006and the imaginary correlation signal 1007. The unified correlationsignal 1011 is provided to a square root device 1012 which takes thesquare root of its input, and generates a final correlation signal 1013.If CSK is used, a maximum correlation pulse 1014 may be achieved onceper symbol code time Ts. The squaring of the correlation pulses causesloss of polarity information in the final correlation signal 1013.

FIG. 15A is a block diagram of another embodiment of a non-coherentspread spectrum receiver using serial correlation for separable real andimaginary parts of the received spread spectrum signal.

The received signal s*(t) 401 is coupled to a dual-integratornon-coherent serial CPM correlator 1102 for recognizing a chip sequencein the received signal s*(t) 401. The dual-integrator non-coherentserial CPM correlator 1102 comprises a first power divider 1103, whichproduces duplicate signals 1104 and 1105, each with a 0-degree phasedelay. A local oscillator 1106 produces a local carrier signal cos ω₁t1107, which is coupled to a second power divider 1108. The second powerdivider 1108 produces duplicate signals, one signal 1109 with a 0-degreephase delay, and another signal 1110 with a 90-degree phase delay.

The signals 1104 and 1109 are coupled to a first multiplier 1111, whichcombines its inputs and produces a first product signal 1112. The firstproduct signal 1112 is coupled to a first low pass filter 1113, whichproduces a first filtered signal 1114 retaining its baseband frequencycomponents.

The signals 1105 and 1110 are coupled to a second multiplier 1116, whichcombines its inputs and produces a second product signal 1117. Thesecond product signal 1117 is coupled to a second low pass filter 1118,which produces a second filtered signal 1119 retaining its basebandfrequency components.

In dual-integrator non-coherent serial CPM correlator 1102, the Qportion of the Real*(t) signal is the same as the I portion of theImag*(t) signal, and the Q portion of the Imag*(t) signal is 180-degreesout of phase (i.e., the inverse) of the I portion of the Real*(t)signal.

First filtered signal 1114 is coupled to a real I multiplier 1121, whichis also coupled to a locally generated signal i(t), that is, the i(t)chip sequence of the correlation sequence (see FIG. 13B). The real Imultiplier 1121 combines its inputs and produces a real I product signal1122.

The first filtered signal 1114 is also coupled to an imaginary Qmultiplier 1123, which is also coupled to a locally generated signal{overscore (q(t))}, that is, the inverted q(t) chip sequence of thecorrelation sequence (see FIG. 13B). The imaginary Q multiplier 1123combines its inputs and produces an imaginary Q product signal 1124.

The second filtered signal 1119 is coupled to an imaginary I multiplier1125, which is also coupled to the locally generated signal i(t). Theimaginary I multiplier 1125 combines its inputs and produces animaginary I product signal 1126.

The second filtered signal 1119 is also coupled to a real Q multiplier1127, which is coupled to a locally generated signal q(t), that is, theq(t) chip sequence of the correlation sequence (see FIG. 13B). The realQ multiplier 1127 combines its inputs and produce a real Q productsignal 1128.

The real I product signal 1122 and the real Q product signal 1128 arecoupled to a real summer 1129, which combines its inputs and produces areal product signal 1130. The imaginary Q product signal 1124 and theimaginary I product signal 1126 are coupled to an imaginary summer 1131,which combines its inputs and produces an imaginary product signal 1132.

The real product signal 1130 is coupled to a real integrator 1133, whichintegrates its input and dumps the sum under control of a control input1134. The real integrator 1133 produces a real correlation signal 1135.

The imaginary product signal 1132 is coupled to an imaginary integrator1136, which integrates its input and dumps the sum under control of acontrol input 1137. The imaginary integrator 1136 produces an imaginarycorrelation signal 1138.

The real correlation signal 1135 is coupled to a real squaring device1139, which computes the square of its input. The imaginary correlationsignal 1138 is coupled to an imaginary squaring device 1140, whichcomputes the square of its input. The two squared values are coupled toa summer 1141, which combines its inputs and produces a unifiedcorrelation signal 1142 which is the sum of the squares of the realcorrelation signal 1135 and the imaginary correlation signal 1136. Theunified correlation signal 1142 is coupled to a square root device 1143,which takes the square root of its input and generates a finalcorrelation signal 1144. The final correlation signal 1144 may have amaximum value once per symbol code time period Ts.

In a particular embodiment, a one-bit quantizer is inserted at theoutput of the first low pass filter 1113 and the second low pass filter1118.

In a preferred embodiment of the FIG. 15A correlator, the real Imultiplier 1121, imaginary Q multiplier 1123, imaginary I multiplier1125, and real Q multiplier 1127 each comprise an inverted XOR gate.Inverted XOR gates are well known in the art; they have a truth table asshown in the table below:

A B Inverted XOR(A,B) −1 −1 +1 −1 +1 −1 +1 −1 −1 +1 +1 +1

In a preferred embodiment, the real summer 1129 and real integrator 1133collectively comprise a multiplexer and integrator. Instead of computingthe individual real I and real Q components, summing them, andintegrating the sum, in a preferred embodiment the individual real I andreal Q components are multiplexed into a single stream and the streamitself integrated.

Likewise, the imaginary summer 1131 and imaginary integrator 1136collectively comprise a multiplexer and integrator. Instead of computingthe individual imaginary I and imaginary Q components, summing them, andintegrating the sum, in a preferred embodiment the individual imaginaryI and imaginary Q components are multiplexed into a single stream andthe stream itself integrated.

In a preferred embodiment, the first squaring device 1139, the secondsquaring device 1140, the summer 1141, and the square root device 1143collectively comprise a device using the Robertson technique forcomputing the square root of the sum of two squares. In the Robertsontechnique, which is known in the art, the norm of a plane vector (thesquare root of the sum of two squares) having coordinates <x,y> may beapproximated as follows:

||<x,y>||=maximum(x,y)+(0.5)minimum(x,y)  (1152)

A preferred embodiment of a Robertson device is shown in FIG. 22 and isdescribed later herein.

FIG. 15B is a block diagram of a spread spectrum receiver usingmulti-bit serial correlation for separable real and imaginary parts ofthe received spread spectrum signal. The FIG. 15B receiver comprises afirst power divider 1153 coupled to a received signal s*(t) 401, a localoscillator 1156, a second power divider 1158, multipliers 1161 and 1166,and low pass filters 1163 and 1168, all of which are similar to the FIG.15A embodiment. Also like the FIG. 15A embodiment, the Q portion of theReal*(t) signal is the same as the I portion of the Imag*(t) signal, andthe Q portion of the Imag*(t) signal is 180-degrees out of phase (i.e.,the inverse) of the I portion of the Real*(t) signal.

The low pass filter 1163 is coupled to a two-bit analog-to-digital (A/D)converter 1164, and the other low pass filter 1168 is coupled to anothertwo-bit A/D converter 1169. The two-bit A/D converters 1164 and 1169each quantize their respective input waveforms, and output a two-bitpattern corresponding to the amplitude of the input waveform. FIG. 15Cis a graph showing a two-bit quantization of an input waveform 1154.Four amplitude regions 1155 are depicted in the graph of FIG. 15C. Whenthe input waveform 1154 (e.g., the output of low pass filter 1163 or1168) is in the highest amplitude region 1155, the A/D converter 1164 or1169 outputs a two-bit pattern I₁I₀ of 11. When the input waveform 1154is in the next highest amplitude region 1155, the A/D converter 1164 or1169 outputs a two-bit pattern I₁I₀ of 10. Likewise, in the next highestamplitude region 1155, the A/D converter 1164 or 1169 outputs a two-bitpattern I₁I₀ of 01, and in the lowest amplitude region 1155 the A/Dconverter 1164 or 1169 outputs a two-bit pattern I₁I₀ of 00.

The inputs of A/D converters 1164, 1169 are sampled once each chipperiod. The outputs 1165, 1170 of the A/D converters 1164, 1169 areprovided to a multi-bit non-coherent serial correlation block 1167. Theoutput 1165 of A/D converter 1164 is coupled to the input of amultiplier 1172, which has another input coupled to a locally generatedi(n) chip signal, which, in a particular embodiment, generates a two'scomplement waveform corresponding to the tri-value return-to-zerowaveform used in FIG. 15A. The output 1165 of A/D converter 1164 is alsocoupled to the input of a second multiplier 1174, which has anotherinput coupled to a locally generated inverse q(n) chip signal, which islikewise a tri-valued signal represented in two's complement format. Theoutput 1170 of A/D converter 1169 is coupled to the input of amultiplier 1171, which has another input coupled to the i(n) chipsignal. The output 1170 of A/D converter 1169 is also coupled to theinput of another multiplier 1173, which has another input coupled to aq(n) chip signal.

Each of multipliers 1171, 1172, 1173 and 1174 is preferably embodied asa digital multiplier that multiplies its inputs and generates a resultin two's-complement format. A preferred input and output truth table foreach of multipliers 1171, 1172, 1173 and 1174 appears in Table 15-1below, wherein i_(c) and q_(c) represent the chip value of the i(t) orq(t) signal at the appropriate time interval. A binary 0-bit for i_(c)or q_(c) represents a −1 chip value, while a binary 1-bit for i_(c) orq_(c) represents a +1 chip value. These values are, as noted for thisparticular embodiment, expressed in two's complement format for thesignals i(n) and q(n).

TABLE 15-1 A/D Output I/Q Signal Result Decimal (I₁ I₀) (i_(c) or q_(c))(O₂ O₁ O₀) Equivalent 0 0 0 0 1 0 +2 0 0 1 1 1 0 −2 0 1 0 0 0 0 +1 0 1 11 1 1 −1 1 0 0 1 1 1 −1 1 0 1 0 0 1 +1 1 1 0 1 1 0 −2 1 1 1 0 1 0 +2

The output from each of multipliers 1171, 1172, 1173 and 1174 comprisesa 3-bit digital signal according to Table 15-1. The outputs frommultipliers 1171, 1172, 1173 and 1174 are coupled to accumulators 1175,1176, 1177, and 1178, respectively. A chip clock signal 1181 isconnected to each of the accumulators 1175, 1176, 1777 and 1178, andcauses the accumulators 1175, 1176, 1177 and 1178 to sample their inputsonce each chip period. Thus, for a symbol code length of 32 chips, theaccumulators 1175, 1176, 1177 and 1178 sample their inputs 32 times fora given symbol code. At each sample time, the accumulators 1175, 1176,1177 and 1178 add the input to a running correlation total. Because theoutputs of A/D converters 1164 and 1169 are represented intwo's-complement notation, the accumulators 1175, 1176, 1177 and 1178effectively carry out addition or subtraction by performing only addingoperations. A dump signal 1182 clears the accumulators at the end ofeach symbol period. For a 32 chip symbol code, the running accumulatortotals will vary between +32 and −32.

Alternatively, instead of using the two's-complement format signals i(n)and q(n), the tri-valued return-to-zero waveforms such as i(t) and q(t)(see FIG. 15A) may be used. In such a case, the accumulators 1175, 1176,1177 and 1178 would each accumulate every other clock cycle in analternating pattern, rather than every clock cycle.

Each accumulator 1175, 1176, 1177, and 1178 outputs a 6-bit digitalaccumulation value. The outputs of accumulators 1176 and 1177 arecoupled to the inputs of a first summer 1179. The outputs ofaccumulators 1175 and 1180 are coupled to the inputs of a second summer1180. Outputs of summers 1179 and 1180 are coupled to a magnitudecalculation block 1185 and a phase calculation block 1187. The magnitudecalculation block 1185 may be embodied as a Robertson device (see, e.g.,FIG. 22). The phase calculation block 1187 may be embodied as, e.g., aphase sector lookup table as shown in and described elsewhere hereinwith respect to FIG. 25B. The magnitude calculation block 1185 and phasecalculation block 1187 output a unified correlation signal 1186 and aphase signal 1188, respectively. The unified correlation signal 1186 maybe, e.g., a 7-bit unsigned digital signal.

Experiment has shown that the correlator of FIG. 15B can realize animprovement in bit error rate (BER) and E_(b)/N_(o) (bit energy/noisedensity) of approximately 1.5 to 2.0 dB over the correlator of FIG. 15A.While two-bit quantization leads to a significant improvement oversingle-bit quantization, it is expected that higher order quantizationwill yield increasingly smaller gains in BER and E_(b)/N_(o) ratio up toa maximum aggregate improvement of about 3 dB. Thus, two-bitquantization provides an advantageous combination of improvedperformance without a large increase in hardware complexity.

FIG. 15D is a block diagram of a portion of another embodiment of aspread spectrum receiver using multi-bit serial correlation forseparable real and imaginary parts of the received spread spectrumsignal. The circuitry shown in FIG. 15D corresponds to the multi-bitnon-coherent serial correlation block 1167 depicted in FIG. 15B, butuses fewer components than the FIG. 15B embodiment.

In FIG. 15D., signal 1165 (see FIG. 15B) and a c(t) chip signal (i.e., acombined i(t) and q(t) signal) are coupled to inputs of a firstmultiplier 1189. Signal 1170 (see FIG. 15B) and the c(t) chip signal arecoupled to inputs of a second multiplier 1190. Multipliers 1189 and 1190each carry out arithmetic operations according to Table 15-1. The outputof the first multiplier 1189 is coupled to the input of a multiplexer1191, and is coupled through an inverter 1193 to the input of anothermultiplexer 1192. The output of the second multiplier 1190 is coupled toan input of each of multiplexers 1191 and 1192.

A multiplexer clock signal 1196 controls the selection of inputs foreach of multiplexers 1191 and 1192. Operation of multiplexer clocksignal 1196 is based on the recognition that the i(t) and q(t) chipsignals are staggered and will be zero every other chip time (see, e.g.,FIG. 13B). The multiplexer clock signal 1196 causes the input of themultiplexers 1191, 1192 to switch so as to ignore the output from themultiplier 1189, 1190 that would be zero because the i(t) or q(t)portion of the c(t) chip signal is zero. Thus, the inputs tomultiplexers 1191, 1192 are switched each chip time.

The output from multiplexer 1191 is input to an accumulator 1194. Theoutput from multiplexer 1192 is input to another accumulator 1195.Accumulators 1194 and 1195 function similarly to accumulators 1175,1176, 1177 or 1178 in FIG. 15B, by performing two's-complementaccumulation of their inputs to keep a running correlation total. Theaccumulators 1194, 1195 are controlled by a chip clock signal 1197 and adump signal 1198, similar to chip clock signal 1181 and dump signal1182, respectively, of FIG. 15B.

The output 1260 of accumulator 1194 is coupled to a magnitudecalculation block 1262 and a phase calculation block 1263. The output1261 of accumulator 1195 is likewise coupled to magnitude calculationblock 1262 and phase calculation block 1263. Magnitude calculation block1262 is similar to magnitude calculation block 1185 of FIG. 15B; phasecalculation block 1187 is likewise similar to phase calculation block1187 of FIG. 15B. Magnitude calculation block 1262 and phase calculationblock 1263 output a unified correlation signal 1264 and a phase signal1265, respectively.

A method of receiving and despreading a spread spectrum signal usingnon-coherent multi-bit serial correlation is also provided. The methodincludes the steps of dividing a spread spectrum signal into first andsecond duplicate signals, demodulating the first signal into areal-I/imaginary-Q signal using a first non-coherent local referencesignal, demodulating the second signal into an imaginary-I/real-Q signalusing a second non-coherent local reference signal having the samefrequency as said first non-coherent local reference signal but phaseoffset therefrom by 90 degrees, converting the real-I/imaginary-Q signalinto a first multi-bit digital signal, converting the imaginary-I/real-Qsignal into a second multi-bit digital signal, correlating the firstmulti-bit digital signal with a chip sequence comprising odd chips andeven chips, accumulating a first correlation total, correlating thesecond multi-bit digital signal with the odd chips and an inverse of theeven chips of the chip sequence, accumulating a second correlationtotal, and combining the first correlation total and the secondcorrelation total to generate a unified correlation output signal.

In one variation of the method, the steps of correlating said firstmulti-bit digital signal, accumulating a first correlation total,correlating said second multi-bit digital signal, accumulating a secondcorrelation total, and combining said first correlation total and saidsecond correlation total comprise the steps of multiplying thereal-I/imaginary-Q signal with said odd chips to generate a real Iproduct signal, multiplying the imaginary-I/real-Q signal with said evenchips to generate a real Q product signal, multiplying theimaginary-I/real-Q signal with said odd chips to generate an imaginary Iproduct signal, multiplying the real-I/imaginary-Q signal with theinverse of said even chips to generate an imaginary Q product signal,individually accumulating at each chip period of said chip sequence thereal I product signal, real Q product signal, imaginary I productsignal, and imaginary Q product signal, summing the accumulated real Iproduct signal and the accumulated real Q product signal into a realcorrelation signal, summing the accumulated imaginary product signal andthe accumulated imaginary Q product signal into an imaginary correlationsignal, and combining the real correlation signal and the imaginarycorrelation signal into a unified correlation signal.

In another variation of the method, the steps of correlating said firstmulti-bit digital signal, accumulating a first correlation total,correlating said second multi-bit digital signal, accumulating a secondcorrelation total, and combining said first correlation total and saidsecond correlation total comprise the steps of multiplying thereal-I/imaginary-Q signal with the chip sequence c(t) to generate areal-I/imaginary-Q product signal, multiplying the imaginary-I/real-Qsignal with the chip sequence c(t) to generate an imaginary-I/real-Qproduct signal, sampling and adding the real-I/imaginary-Q productsignal into a first running correlation total (e.g., a real correlationtotal) the imaginary-I/real-Q product signal into a second runningcorrelation total (e.g., an imaginary correlation total) for the oddchips of the chip sequence, and sampling and adding theimaginary-I/real-Q product signal into said first running correlationtotal an inverse of the real-I/imaginary-Q product signal into a secondrunning correlation total for the even chips of the chip sequence.

FIG. 16 shows a block diagram of a first spread spectrum receiver usingself-synchronized correlation for separable real and imaginary parts ofthe received spread spectrum signal.

The received signal s*(t) 401 is coupled to a self-synchronized CPMcorrelator 1202 for recognizing a correlation sequence in the receivedsignal s*(t) 401. The self-synchronized CPM correlator 1202 comprises apower divider 1203, which produces duplicate signals, Real*(t) 1204having a 0-degree phase delay, and Imag*(t) 1205 having a 90-degreephase delay. Real*(t) 1204 and Imag*(t) 1205 are the real and imaginaryparts of the received signal s*(t) 401.

The Real*(t) signal 1204 is coupled to a real correlator 1206, whichdivides its input signal by a power divider (not shown) or othersuitable means. The real correlator 1206 comprises a real I multiplier1207, which is also coupled to a local carrier signal cos ω₁t. The realI multiplier combines its inputs and produces a real I product 1208. Thereal I product 1208 is coupled to a real I low pass filter 1209, whichfilters its input and produces a filtered real I signal 1210.

The filtered real I signal 1210 is coupled to a real Iself-synchronizing correlator 1211, such as a correlator usingself-synchronizing techniques described in application Ser. No.08/432,913 entitled “Method and Apparatus for Despreading SpreadSpectrum Signals,” filed May 1, 1995 in the name of inventors RobertGold and Robert C. Dixon, which application is assigned to the assigneeof the present invention and hereby incorporated by reference.

The real I self-synchronizing correlator 1211 comprises a shift register1212 having a plurality of chips 1213 and a plurality of taps 1214coupled to selected chips 1213. The taps 1214 are coupled to a first tapmultiplier 1215, which combines its inputs to produce a product which isthereafter coupled to a second tap multiplier 1216. The second tapmultiplier 1216 is also coupled to the filtered real I signal 1210. Thesecond tap multiplier 1216 combines its inputs and produces a real Icorrelation signal 1217.

The real correlator 1206 further comprises a real Q multiplier 1218,which is coupled to a local carrier signal sin ω₁t. The real Qmultiplier 1218 combines its inputs and produces a real Q product 1219.The real Q product 1219 is coupled to a real Q low pass filter 1220,which filters its input and produces a filtered real Q signal 1221.

The filtered real Q signal 1221 is coupled to a real Qself-synchronizing correlator 1222, which produces a real Q correlationsignal 1223.

The Imag*(t) signal 1205 is coupled to an imaginary correlator 1224,which divides its input signal by a power divider (not shown) or othersuitable means. The imaginary correlator 1224 comprises an imaginary Imultiplier 1244, which is also coupled to a local carrier signal cosω₁t. The imaginary I multiplier 1244 combines its input and produces animaginary I product 1225. The imaginary I product 1225 is coupled to animaginary I low pass filter 1226, which filters its input and produces afiltered imaginary I signal 1227.

The filtered imaginary I signal 1227 is coupled to an imaginary Iself-synchronizing correlator 1228, which produces an imaginary Icorrelation signal 1229.

The imaginary correlator 1224 comprises an imaginary Q multiplier 1230,which is also coupled to a local carrier signal sin ω₁t. The imaginary Qmultiplier 1230 combines its inputs and produces an imaginary Q product1231. The imaginary Q product 1231 is coupled to an imaginary Q low passfilter 1232, which filters its input and produces a filtered imaginary Qsignal 1233.

The filtered imaginary Q signal 1233 is coupled to an imaginary Qself-synchronizing correlator 1234, which produces an imaginary Qcorrelation signal 1235.

The real I correlation signal 1217 and the imaginary I correlationsignal 1229 are coupled to squaring devices 1236 and 1237 respectively,the outputs of which are coupled to a summer 1238, to produce a unifiedI correlation signal 1239. The unified I correlation signal 1239 iscoupled to a square root device 1250 which takes the square root of itsinput and generates an final I correlation signal 1251.

The real Q correlation signal 1223 and the imaginary Q correlationsignal 1235 are coupled to squaring devices 1240 and 1241 respectively,the outputs of which are coupled to a summer 1242, to produce a unifiedQ correlation signal 1243. The unified Q correlation signal 1243 iscoupled to a square root device 1252 which takes the square root of itsinput and generates an final Q correlation signal 1253.

Embodiments and other aspects of the inventions described herein,including the system embodiments described below, may be made or used inconjunction with inventions described, in whole or in part, in thepatents, publications, or copending applications referred to herein aswell as in copending U.S. Pat. No. 5,455,822, entitled “Method andApparatus for Establishing Spread Spectrum Communication,” or copendingU.S. patent application Ser. No. 08/284,053, filed Aug. 1, 1994, ” bothof which applications are hereby incorporated by reference as if setforth fully herein.

FIG. 17A is a block diagram of a preferred transmitter.

In a preferred embodiment, a spread spectrum transmitter 1337 operatesin a cellular environment like that described with respect to FIG. 2.The transmitter 1337 may be associated with either a base station or auser station in such a cellular environment. In a preferred embodiment,the transmitter 1337 operates according to an over-air protocol forcommunication between the base station and the user station, in whichtransmission is time-division duplex between the base station and theuser station in a single frame, and is time-division multiplexed amongmultiple user stations in a repeated pattern of frames. Other andfurther details regarding a preferred over-air communication protocalmay be found in U.S. Pat. No. 5,455,822 application Ser. No. 08/284,053cited above. However, the present invention will work in a variety ofdifferent communication environments, cellular or otherwise, andaccording to a variety of different protocols, whether or not suchprotocols make use of time-division duplexing or time-divisionmultiplexing.

A preferred communication protocol is depicted in FIG. 17D. As shown inFIG. 17D, a polling loop 1380 (“major frame”) comprises a plurality oftime slots 1381 (“minor frames”). Each minor frame 1381 preferablycomprises communication between a base station (e.g., cellular station)and a user station (e.g., mobile user) in time division duplex—that is,the base station transmits to a user station and the user stationtransmits back to the base station within the same minor frame 1381.

More specifically, as shown in an exploded view in FIG. 17D, a minorframe 1381 preferably comprises a mobile or user transmission 1382preceding a base transmission 1383. The minor frame 1381 also comprisesa variable radio delay gap 1384 preceding the user transmission 1382,followed by a turn-around gap 1388 and a guard time gap 1389. After gap1389 is the base transmission 1383, which is followed by anotherturn-around gap 1393. The user transmission 1382 comprises a preamble1385, a preamble sounding gap 1386, and a user message interval 1387.The base transmission comprises a preamble 1390, a preamble sounding gap1391, and a base message interval 1392.

Another communication protocol is shown in FIG. 17B. While operation ofthe transmitter in FIG. 17A is generally described with reference to theFIG. 17B protocol, the same techniques may be applied for use with thepreferred protocol shown in FIG. 17D. In the particular protocol of FIG.17B, a polling loop 1301 (“major frame”) comprises a plurality of timeslots 1302 (“minor frames”). Each minor frame 1302 comprisescommunication between a base station (e.g., cellular station) and a userstation (e.g., mobile user) in time division duplex—that is, the basestation transmits to a user station and the user station transmits backto the base station within the same minor frame 1302.

More specifically, as shown in an exploded view in FIG. 17B, a minorframe 1302 comprises a power control pulse transmission 1304 from theuser station to the base station, a base station transmission 1305, anda user station transmission 1306, each of which is surrounded by guardbands 1303. Details regarding the power control pulse transmission 1304may be found in application Ser. No. 08/284,053, filed Aug. 1, 1994, andincorporated herein by reference. The base station transmission 1305 andthe user station transmission 1306 have a similar structure; thus, thefollowing description regarding the base station transmission 1305applies equally to the user station transmission 1306.

The base station transmission 1305 comprises an interframe gap 1351, amatched filter code 1352, a first fill code 1353, a data sequence 1354,and a second fill code 1355 similar to the first fill code 1353. Theinterframe gap 1351 may be four chips in duration; the matched filtercode 1352 may be 48 chips duration; the first fill code 1353 may be 16chips in duration; the data sequence 1354 may be comprised of one ormore symbol codes, each of which may be 32 chips, 128 chips, 2048 chips,or some other number of chips in duration depending upon a data rate fortransmission between the base station and the user station; and thesecond fill code 1355 may be a sufficient number of chips in duration tocomplete the minor frame 1302. A plurality of minor frames 1302 maycomprise a channel.

The fill codes 1353, 1355 preferably each comprise a code that has a lowcross-correlation with each of the symbol codes, and may form a repeatedpattern such as “0 1 0 1 . . . ” or “0 0 1 1 . . . . ”0 The interframegap 1351 may have the same code as one or both of the fill codes 1353,1355. Fill codes 1353, 1355 are generated primarily for the purpose ofstarting the modulator in a known state at the beginning of atransmission, and to avoid having to turn the transmitter off and on forthe time period while the fill code 1305 is transmitted. The fill codes1353, 1355 may further be selected to improve the spectralcharacteristics of the overall transmission.

The transmitter 1337 of FIG. 17A is a preferred means for generating abase station transmission 1305 (or 1387 of FIG. 17D) or a user stationtransmission 1306 (or 1392 of FIG. 17D) using CPM techniques such asthose described elsewhere herein. A serial data stream 1321 ofinformation to be transmitted is provided to the transmitter 1337 andconverted to parallel data by a serial-to-parallel shift register 1322.The parallel data output by the serial-to-parallel shift register 1322is used to select from among a plurality of symbol codes stored in asymbol code table 1323. Each symbol code, as mentioned, is preferably 32chips in length and represents a predetermined number of data bits(preferably 5 data bits) from the serial data stream 1321.

In addition to storing various symbol codes in the symbol code table1323, the transmitter also comprises a matched filter code generator1324 capable of generating a matched filter code 1352, and a fill codegenerator 1325 (which may be a table) capable of generating fill codes1353, 1355. The symbol code table 1323, matched filter code generator1324, and fill code generator 1325 are selectively accessed by a controlcircuit 1320 for constructing a transmission such as a base stationtransmission 1305 or user station transmission 1306. A transmission maybe constructed, for example, by concatenating or appending consecutivesymbol codes, fill codes, and other code sequences as necessary togenerate the appropriate chip sequence. Although connections are notexpressly shown, the control circuit 1320 has control outputs 1339connected to various parts of the circuit for the purpose of exercisingsynchronous control.

In a preferred embodiment, timing information is generated with a clockcircuit 1307 such as a crystal oscillator. The clock circuit 1307produces a 20 megahertz (MHz) clock signal and is coupled to an input ofa clock chain 1308. The clock chain 1308 generates a plurality of outputclock signals in a manner known in the art. The clock chain 1308 has asoutputs a 20 MHz clock signal 1309, a 10 MHz clock signal 1310, a 5 MHzclock signal 1311, and a 2.5 MHz clock signal 1312.

In a preferred embodiment, the 5 MHz clock signal 1311 is coupled to aloop counter 1313, which, among other things, counts chips over thecourse of each minor frame 1302. The loop counter 1313 produces a chipcount signal 1314, a symbol count signal 1315, and a channel countsignal 1316. The channel or loop count signal 1316 indicates which minorframe 1302 is active within the polling loop 1301. Thus, if there are 32minor frames 1302 in a polling loop 1301, the channel count signal 1316counts from 0 to 31 and then resets. When the channel count signal 1316indicates an active minor frame 1302 in which the transmitter 1337 isauthorized to transmit, the control circuit 1320 may issue commands totransmit information at the appropriate time.

The symbol count signal 1315 keeps track of how many symbols have beentransmitted by the transmitter 1337 in the data sequence 1354. Thus, ifthe transmitter is to transmit 16 consecutive symbols as part of thedata sequence 1354, then the symbol count signal 1315 counts from 0 to15 and then resets.

The chip count signal 1314 keeps track of how many chips have beentransmitted by the transmitter 1337 for the current symbol in the datasequence 1354. Thus, if each symbol code is 32 chips in length, the chipcount signal 1314 counts from 0 to 31 and then resets. The chip countsignal 1314 also provides timing information for those circuits in thetransmitter which are clocked at each chip time Tc.

The chip count signal 1314, the symbol count signal 1315, and thechannel count signal 1316 are coupled to a state decoder 1317, whichdetermines whether the current chip is part of the matched filter code1352, the fill code 1305, or a data sequence symbol code 1306, and whichgenerates a selection signal 1318 and a set of control signals 1319. Thecontrol signals 1319 are coupled to a control circuit 1320.

As mentioned, a serial data stream 1321 of data to be transmitted iscoupled to a serial-to-parallel shift register 1322, which converts theserial data stream 1321 to a sequence of 5-bit parallel symbols. Thesequence of symbols is coupled to an input of a symbol code table 1323,which selects for each symbol a specific symbol code unique to thesymbol.

The chip count signal 1314 is coupled to the symbol code table 1323, thematched filter code generator 1324, and the fill code generator 1325.Outputs of the symbol code table 1323, the matched filter code generator1324, and the fill code generator 1325 are coupled to inputs of a 3-1multiplexer 1326. A control input of the 3-1 multiplexer 1326 is coupledto the selection signal 1318 from the control circuit 1320. The 3-1multiplexer 1326 thus generates an output chip stream 1327 in accordancewith the commands provided by the control circuit 1320. Specifically,the control circuit 1320 may select a fill code to fill the interframegap 1351 from the fill code generator 1325, a matched filter code 1352from the matched filter code generator 1324, a first fill code 1353 fromthe fill code generator 1325, one or more symbol codes (depending on theamount of data to be transmitted and the data rate) corresponding to thedata sequence 1354 from the symbol code table 1323, and a second fillcode 1355 from the fill code generator 1325, in order to construct atransmission such as a base station transmission 1305 or a user stationtransmission 1306.

The output chip stream 1327 is coupled to a demultiplexer 1328, whichseparates its input chip stream into an I chip stream 1329 and a Q chipstream 1330, under control of the 2.5 MHz clock signal 1312 (i.e., thedemultiplexer 1328 is clocked at half the chip rate Rc). The I chipstream 1329 and the Q chip stream 1330 are connected to a waveformgenerator 1338 which generally constructs appropriate output waveformsbased on the contents of the I chip stream 1329 and the Q chip stream1330.

The waveform generator 1338 comprises an I lookup table 1332 and a Qlookup table 1334, each of which comprises memory such as ROM. The Ilookup table 1332 and the Q lookup table 1334 each contain fifteendigitized values for amplitude outputs of the P(t) devices 305 (for I)and 306 (for Q) shown in FIG. 6. Thus, by changing the contents of thelookup tables 1332, 1334 appropriately, the shape of the outputwaveforms may be suitably altered, allowing transmission of MSK, SQAM,GMSK, SQORC, or other similar format as desired.

The I lookup table 1332 receives as its inputs both the present I chipfrom the I chip stream 1329 and the previous I chip from the I chipstream 1329 as stored in an I delay element 1331 (e.g., a latch). Byhaving available the immediate past I chip and the present I chip, thetransmitter knows what type of transition is occurring in the I chipstream 1329—that is, whether the I chip stream 1329 is undergoing a 0/0transition, a 0/1 transition, a 1/0 transition, or a 1/1 transition. Thetype of transition determines the shape of the output waveform. The Ilookup table 1332 provides as output eight sequential I waveformcommands or “samples” per I chip time (i.e., 2Tc) which are connected toa digital-to-analog converter (DAC) for constructing a suitablewaveform. The I lookup table 1332 is provided a clock input of 20 MHz sothat eight I waveform commands may be output per I chip time. In thetransmitter 1337 shown in FIG. 17, the DAC for the I chip stream 1329comprises a 4-15 decoder 1335 which selects one of 15 possible outputlines, coupled to a resistor ladder (not shown) and a low pass filter(not shown). Of course, other types of DAC's would be suitable for thispurpose.

Table 17-1 below shows an example of how the 15 outputs of the 4-15decoder 1335 relate to specific voltages to be output by the DAC tocreate a SQAM waveform varying between 1.5 V and 3.5 V:

TABLE 17-1 Decoder (Hex) Output Amplitude (V) 0 1.5 1 1.5674 2 1.700 31.8414 4 1.900 5 3.100 6 3.1586 7 3.300 8 3.4326 9 3.500 A 3.2071 B2.8827 C 2.500 D 2.1173 E 1.7929

Table 17-2 below shows a sequence of eight selected values according toTable 17-1 for constructing an appropriate waveform depending on whattype of transition is occurring in the I chip stream:

TABLE 17-2 Transition Decoder Output Sequence 0 -> 0 0, 1, 2, 3, 4, 3,2, 1 0 -> 1 0, 1, E, D, C, B, A, 8 1 -> 0 9, 8, A, B, C, D, E, 1 1 -> 19, 8, 7, 6, 5, 6, 7, 8

An output corresponding to the Q chip stream 1330 is generated in asimilar manner to that of the I chip stream 1329. The Q lookup table1334 receives as its inputs both the present Q chip from the Q chipstream 1330 and the previous Q chip from the Q chip stream 1330 asstored in a Q delay element 1333. Based on its inputs, the Q lookuptable 1334 determines what type of transition is occurring in the Q chipstream 1330. An output of the Q lookup table 1334 is coupled to a 4-15decoder 1336, which selects one of 15 output lines for sending a signalto a DAC configured in a similar manner to that described with respectto the I chip stream 1329.

Thus, the contents of the I lookup table 1332 and the Q lookup table1334 are selected to generate an i(t) output waveform and a q(t) outputwaveform, respectively. An example of an output SQAM waveform 1370according to the technique described above and the values set forth inTables 17-1 and 17-2 is shown in FIG. 17C. The waveform 1370 comprises a0/0 transition 1372, a 0/1 transition 1373, and a 1/1 transition 1374.Each transition 1372, 1373, 1374 comprises eight discrete points 1371corresponding to values selected by the 4-15 I lookup table 1332 (or Qlookup table 1334). The effect of the low pass filter (not shown) at theoutput of the waveform generator 1338 smooths the shape of the waveform1370 between discrete points 1371.

Table 17-3 shows an illustrative matched filter code 1352. In apresently preferred embodiment, the matched filter lo code generator1324 is configured to generate the code shown below in Table 17-3.

TABLE 17-3 Hexadecimal Value Binary Value 40 01000000 3E 00111110 3400110100 B3 10110011 1A 00011010 A6 10100110

Selection of a matched filter code 1352 for a particular applicationdepends on the symbol codes (in a CSK system) or other chip codes beingused; generally, the matched filter code 1352 is selected for low crosscorrelation with the other chip codes used in the particularcommunication environment.

Table 17-4 shows a presently preferred set of 32 symbol codes. In apreferred embodiment, the symbol code table 1323 along with theappropriate commands from the control circuit 1320 are configured togenerate a sequence of symbol codes selected from the set of 32 symbolcodes shown in Table 17-4, in response to a sequence of 5-bit parallelsymbols.

TABLE 17-4 Symbol Symbol Code (Hex) Symbol Symbol Code (Hex) 00 0544D65E10 0E4424A1 01 5B118E0B 11 5B1171F4 02 3D77E66D 12 3D771792 03 6822BD3613 682242C7 04 014BD451 14 014B2BAE 05 541E8104 15 541E7EFB 06 3278E76216 3278189D 07 672DB237 17 672D4DC8 08 0EBBDBA1 18 0EBB245E 09 5BEE8EF419 5BEE710B 0A 3D88E892 1A 3D86176D 0B 68DDBDC7 1B 68DD4238 0C 01B4D4AE1C 01B42B51 0D 54E181FB 1D 54E17ED4 0E 3287E79D 1E 32671862 0F 67D2B2C81F 67D24D37

FIGS. 18, 19, 21A and 21B collectively illustrate a preferred receiver.

The illustrated receiver generally operates by correlating to apreceding spread spectrum code (e.g., matched filter code 1352 of FIG.17B, or preamble 1385 or 1390 shown in FIG. 17D) with a non-coherentparallel correlator (such as the two-register non-coherent CPMcorrelator 802 depicted in FIG. 12) to achieve synchronization for aplurality of serial correlators (such as the dual-integratornon-coherent serial CPM correlators 1102 depicted in FIG. 15A). Theserial CPM correlators are then used for correlating to a followingmessage (e.g., data sequence 1354 of FIG. 17B, or messages contained inthe user message interval 1387 or base message interval 1392 of FIG.17D). However, many alternate configurations using, for example, onlyparallel correlators, only serial correlators, or various combinationsof parallel and serial correlators, may be used in the receiver withoutdeparting from the scope and spirit of the invention. In a preferredembodiment, the multi-bit serial correlators of FIG. 15B or 15D are usedfor the plurality of serial correlators.

A preferred embodiment of a receiver is shown in part in FIGS. 21A and21B. Standard electrical engineering symbols and terms are used in FIGS.21A and 21B; thus, the following explanation will be limited to relatingFIGS. 21A and 21B to the prior description of the invention in some itsvarious embodiments.

A received signal 2001 is provided to an IF amplifier shown in FIG. 21A.The received signal 2001 may undergo prior conditioning and may bedownconverted to an intermediate frequency for processing. The receivedsignal 2001 is coupled to a capacitor C4 which passes the high frequencycomponents of the received signal 2001. The output of the capacitor C4is coupled to a first integrated chip Ul which is preferably an MC13155chip manufactured by Motorola. Specifically, the output of capacitor C4is coupled to a hardlimit amplifier 2003 located on the first integratedchip U1 for hardlimiting the output of capacitor C4. The hardlimitamplifier provides a differential output comprising a first differentialoutput signal 2010 and a second differential output signal 2011.

The differential output signals 2010, 2011 are coupled to a secondintegrated circuit U2 which, as shown in FIG. 21B, is preferably anRF2701 chip manufactured by RF Micro Devices. Specifically, thedifferential output signals 2010, 2011 are coupled to a differentialamplifier 2033 which produces an amplified output signal 2030. Theamplified output signal 2030 is split into two branches by a powerdivider (not shown) and coupled via a first branch to a first multiplier2031 (e.g., multiplier 1111 in FIG. 15A) and via a second branch to asecond multiplier 2032 (e.g., multiplier 1116 in FIG. 15A). The firstmultiplier 2031 has as a second input a reference signal 2036 comprisinga first square wave of frequency ω₁t (which, after low pass filtering,becomes cos ω₁t), and the second multiplier 2032 has as another input areference signal 2037 comprising a second square wave of frequency ω₁t(which, after low pass filtering, becomes sin ω₁t) phase offset from thefirst square wave by 90 degrees.

The reference signals 2036, 2037 are generated from a local oscillator(not shown) which provides a local oscillator signal 2025 to filtercapacitor C39, the output of which is connected to the second integratedchip U2. Specifically, the output of capacitor C39 is connected to anamplifier 2038, the output of which is coupled to a quad divide-by-twocircuit 2039 for splitting its input into two reference signals 2036,2037, the first reference signal 2036 having a 0-degree delay and thesecond reference signal 2037 having a 90-degree delay. The outputs ofmultipliers 2031 and 2032 are amplified by a first output amplifier 2034and a second output amplifier 2035, respectively.

The output of the first output amplifier 2034 is coupled to a first lowpass filter 2023, and the output of the second output amplifier 2035 iscoupled to a second low pass filter 2024. The output of the first lowpass filter 2023 is connected to one input of a first comparator 2027.The output of the second low pass filter 2024 is connected to one inputof a second comparator 2040. The first comparator 2027 and secondcomparator 2040 each have as a second input a DC threshold signal 2041generated by a DC bias circuit 2022. The DC threshold signal 2041 iscoupled to the first comparator 2027 by a low pass filter comprisingcapacitor C52 and resistor R36, and similarly to the second comparator2040 by a low pass filter comprising capacitor C53 and resistor R37. Thefirst comparator 2027 and second comparator 2040 provide output signals2028 and 2029, respectively, each of which comprises a TTL level signalsuitable for further processing using digital circuits. In particular,output signals 2028 and 2029 may each comprise a square wave signalhaving values of +1 and 0 times a fixed voltage.

In a preferred embodiment, the output signals 2028 and 2029 are sampledand provided to remaining circuitry as shown in FIGS. 18 and 19.Specifically, the output signals 2028 and 2029 are sampled twice perchip time (i.e., at 10 MHz) as provided to the circuitry in FIG. 18, andonce per chip time (i.e., at 5 MHz) as provided to the circuitry in FIG.19.

FIG. 18 is a block diagram of a noncoherent matched filter andassociated receiver components.

In a preferred embodiment, a digitally sampled version of a real portionand an imaginary portion of the received signal s*(t) 401 are input tothe circuitry of FIG. 18. Thus, a real-I/imaginary-Q signal 1401 isconnected to signal 2028 shown in FIG. 21B, and input to an even/oddshift register 1402. An imaginary-I/real-Q signal 1451 is connected tosignal 2029 shown in FIG. 21B, and input to an even/odd shift register1452.

In the FIG. 18 embodiment, the even/odd shift register 1402 is 96 bitslong. Because the real-I/imaginary-Q signal 1401 is clocked at twice thesystem clock rate, every other odd chip (rather than every odd chip) ofthe even/odd shift register 1402 is selected and compared with the oddchips of the matched filter code 1403. In a preferred embodiment,matches between every other odd chip of the even/odd shift register 1402and the odd chips of the matched filter code are compared. The chipmatches are coupled to real adder 1404 for counting. Every other evenchip (rather than every even chip) of the even/odd shift register 1402is compared with the even chips of the matched filter code 1403, and theresult of the comparison coupled to the imaginary adder 1405 forcounting.

In the FIG. 18 embodiment, the even/odd shift register 1452 is 96 bitslong. Every other odd chip of the even/odd shift register 1452 iscompared with the odd chips of the matched filter code 1403. Matchesbetween every other odd chip of the even/odd shift register 1452 and theodd chips of the matched filter code are compared. The chip matches arecoupled to the real adder 1404 for counting. Every other even chip ofthe even/odd shift register 1452 is compared with the even chips of thematched filter code 1403, and coupled to the imaginary adder 1405 forcounting.

Although the FIG. 18 embodiment is configured to receive a preamble 48chips in length, in a preferred embodiment the FIG. 18 receiver isconfigured to receive a preamble of 128 chips in length, in accordancewith the preferred FIG. 17D message format. In this latter embodiment,the even/odd shift register 1402 and the odd/even register 1452 are each256 bits long, and the related circuitry is scaled up appropriately.

In the FIG. 18 embodiment, the real adder 1404 has 24 individual bitinputs, each one of which may be a logical “0” to indicate no match or alogical “1” to indicate a match. The real adder 1404 generates a 5-bitreal sum 1406, which represents the absolute value of the number of oddchips that were matched. The imaginary adder 1405 has 24 individual bitinputs and generates a 5-bit imaginary sum 1407 representing theabsolute value of the number of even chips that were matched.

The real sum 1406 and the imaginary sum 1407 are coupled to a Robertsondevice 1408, which computes an approximation of a square root of the sumof the squares of the real sum 1406 and the imaginary sum 1407, asdescribed herein.

An output of the Robertson device 1408 is coupled to an input of acomparator 1409, which compares the output of the Robertson device 1408with a threshold value 1410. In a preferred embodiment, the thresholdvalue is preset, or may be set in response to a control on the receiver.The threshold value may also be set in a variety of other manners, suchas in response to a control in the transmission or to receivingconditions.

A comparator 1409 generates an output pulse 1411. The output pulse is alogical “1” when the input 1430 exceeds the threshold 1410, and alogical “0” when it does not. The output pulse 1411 may have a durationof 100, 200, 300 or 400 nanoseconds.

The output pulse 1411 is coupled to an input of a center seekingdetector circuit 1412. The center seeking detector circuit 1412 receivesthe output pulse 1411 and generates a set clock pulse 1413 which denotesthe end of the received matched filter code 1352, and which is alignedwith the center of a received chip so that the receiver clock can besynchronized with the center of each received chip in a received chipstream.

In a preferred embodiment, the center seeking detector circuit 1412counts the number of logic “1” values in the output pulse 1411 (i.e.,the length of time that the output of the Robertson device 1408 exceedsthe threshold value 1410), thereby measuring the duration of the outputpulse 1411 (e.g., from 1 to 4 clock periods of the 10 MHz clock,corresponding to up to four bits of the even/odd shift register 1402 andthe even/odd shift register 1452). The center seeking detector circuit1412 generates a set clock pulse 1413 which re-initializes a systemclock for serial correlation by a set of serial correlators (see FIG.19) after a preset delay period. The preset delay period ensures thatthe serial correlation clock is properly synchronized with the center ofthe output pulse 1411. Preferred delay periods are shown in Table 18-1:

TABLE 18-1 Length of Output Pulse Delay in Nanoseconds 1 50 2 100 3 1504 200

The system clock may be re-initialized at the start of each minor frame1302.

The set clock pulse 1413 is coupled to a clock chain 1415, which is alsocoupled to a locally generated 40 MHz clock signal 1416. The clock chain1415 generates a 20 MHz clock signal 1417, a 10 MHz clock signal 1418,and a 5 MHz clock signal 1419. In a preferred embodiment, the 5 MHzclock signal 1419 is coupled to, among other things, a set of 32 serialcorrelators (see FIG. 19).

The 5 MHz clock signal 1419 is coupled to a loop counter 1420. The loopcounter 1420 counts the number of chips received and generates a chipcount signal 1421, a symbol count signal 1422, and a channel or loopcount signal 1423, similar to the chip count signal 1314, symbol countsignal 1315, and channel count signal 1316, respectively, generated inthe transmitter 1337.

The chip count signal 1421, symbol count signal 1422, and channel countsignal 1423 are coupled to a state decoder 1424, which determineswhether the received chip is part of the matched filter code 1352, thefill code 1305, or a data sequence symbol code 1306, similar to thestate decoder 1317 in the transmitter 1337, and generates a stateidentifier 1425, similar to the selection signal 1318 generated in thetransmitter 1337. The state identifier 1425 is coupled to an input ofthe center seeking detector circuit 1412.

The state decoder 1424 generates a synchronization signal 1426, which iscoupled to a set of 32 serial correlators (see FIG. 19). The statedecoder 1424 also generates a plurality of control signals 1427, whichare coupled to a control circuit 1428. Although connections are notexpressly shown, the control circuit 1428 has control outputs 1429connected to various parts of the circuit for the purpose of exercisingsynchronous control.

The center seeking detector circuit 1412 also generates a set statesignal 1414 which may be used to place the loop counter 1420 in a knownstate, or to reset the individual count signals 1421, 1422 and 1423associated with the loop counter 1420.

Operation of the center seeking circuit 1412 in relation to the otherelements shown in FIG. 18 may be further explained with reference toFIG. 20, which is a diagram of a series of correlation pulses 2007,2011, 2012, 2013 and 2014 corresponding to output pulse 1411 over aseries of minor frames 1302. A first correlation pulse 2007 is detectedas shown in FIG. 18. The first correlation pulse 2007 has a duration ofthree sample periods 2008. Thus, according to Table 18-1, the centerseeking circuit 1412 generates a set clock pulse 1413 having a delay of150 nanoseconds.

The control circuit 1428 determines, based in part on the count signals1421 through 1423 of the loop counter 1420, the next minor frame 1302 inwhich the receiver is to be active. In many cases, a receiver receivesin only one minor frame 1302 per major frame 1301 located in the samerelative position chronologically from major frame 1301 to major frame1301. Thus, in the next active minor frame 1302, the receiver opens atiming window 2010 during which the next output pulse 1411 is expected.The timing window may be, for example, 1.6 milliseconds in duration, andwill be opened a predetermined time length 2009 before the next outputpulse 1411 is expected assuming no deviation between the transmitter andreceiver clocks between transmissions.

In the example of FIG. 20, a second correlation pulse 2011 is generatedduring the timing window 2010 but some amount of time after expected.The second correlation pulse 2011 is two sample periods in duration, andthus, according to Table 18-1, the center seeking circuit 1412 generatesa set clock pulse 1413 having a delay of 100 nanoseconds. In thefollowing active minor frame 1302, the timing window 2010 has beenshifted in relative time based upon the second correlation pulse 2011,and a third correlation pulse 2012 is generated within the timing window2010 but some amount of time before expected. The third correlationpulse 2012 is four sample periods in duration and causes a set clockpulse 1413 having a delay of 200 nanoseconds.

Similarly, a fourth correlation pulse 2013 and fifth correlation pulse2014 are generated in the next two active minor frames 1302. However, inthe next active minor frame 1302 no correlation pulse is generated;thus, the receiver remains inactive because synchronization has not beenachieved. Measures may be undertaken at such a point to reacquiresynchronization and/or re-establish proper timing.

FIG. 19 is a block diagram of a preferred system of serial correlatorsoperating in parallel with one another and operating in conjunction withthe circuitry of FIG. 18 and FIGS. 21A and 21B.

A digitally sampled version of a real portion and an imaginary portionof the received signal s*(t) 401 are input to the circuitry of FIG. 19.Thus, a real-I/imaginary-Q signal 1511 and an imaginary-I/real-Q signal1512 are generated from the received signal s*(t) 401.

In a preferred embodiment, the 5 MHz clock signal 1419 and thesynchronization signal 1426 as described in FIG. 18 are coupled to acount chain 1501, which generates an output synchronization signal 1502for the serial correlators and a counter clock 1503.

The 5 MHz clock signal 1419, the synchronization signal 1502, thecounter clock 1503, the real-I/imaginary-Q signal 1511 and theimaginary-I/real-Q signal 1512 are each coupled to a set of 32 serialcorrelators 1504. A set of 32 symbol generators 1505, one for eachsymbol 00 through 1F (hexadecimal), are also coupled to each serialcorrelator 1504.

Each serial correlator 1504 recognizes a single one of the 32 symbolcodes and generates a magnitude signal 1506 indicating the number ofagreements with that symbol code. The 32 magnitude signals 1506 arecoupled to a best-of-M device 1507, which determines which one of the 32magnitude signals 1506 has the greatest value and generates an outputsymbol 1508 based thereon. If serial output data is desired, the outputsymbol 1508 may be coupled to a parallel-to-serial shift register 1509,which generates a sequence of serial data bits 1510 in response.

An exploded view of an individual serial correlator 1504 is also shownin FIG. 19. The serial correlator 1504 shown in the FIG. 19 embodimentoperates in a conceptually similar manner to the dual-integratornon-coherent serial CPM correlator 1102 depicted in FIG. 15A. In analternative preferred embodiment, the 32 serial correlators operateaccording to the correlator embodiments described with respect to FIG.15B or 15D.

In a preferred embodiment, the real-I/imaginary-Q signal 1511 is coupledto XNOR gates 1551 and 1552, and the imaginary-I/real-Q signal 1512 iscoupled to XNOR gate 1552. XNOR gates generate the inverted XOR of theirinputs. The XNOR gates 1551 and 1552 perform the function of multipliers1121, 1123, 1125 and 1127 depicted in FIG. 15A. Each serial correlator1504 is programmed to correlate to a different symbol code; accordingly,the appropriate symbol code is clocked into the XNOR gates 1551, 1552and 1554 from the symbol generator 1505. The symbol code is inverted byinvertor 1553 before being received by XNOR gate 1554 because XNOR gate1554 operates on the inverse of the q(t) signal.

Summation and integration is carried out by a pair of multiplexers 1555,1556 and counters 1557, 1558. The outputs of the XNOR gates 1551 and1552 are coupled to an real multiplexer 1555; the outputs of the XNORgates 1552 and 1554 are coupled to an imaginary multiplexer 1556. Thecounter clock 1503 is coupled to a control input of the real multiplexer1555 and the imaginary multiplexer 1556 in order to control theintegrate-and-dump function. The outputs of the real multiplexer 1555and the imaginary multiplexer 1556 are coupled to the enable inputs ofthe real counter 1557 and the imaginary counter 1558, respectively.Because the received I and Q signals are time staggered, the realmultiplexer 1555 selects between real I and real Q signals and providesthem to the real counter 1557 to effectively sum and integrate the realI and real Q signals; the imaginary multiplexer 1556 and imaginarycounter 1558 operate in an analogous manner with respect to imaginary Iand imaginary Q signals.

A reset command may be provided to the real counter 1557 and theimaginary counter 1558 to perform an operation analogous to a “dump” aswould be carried out with integrate-and-dump circuits shown in FIG. 15A.

The output of the real counter 1557 and of the imaginary counter 1558are coupled to a Robertson device 1559, which computes an approximationto the root of the sum of the squares of its inputs. An output of theRobertson device 1559 is output from the serial correlator 1504, andgenerally corresponds to the final correlation signal 1144 such asdescribed with respect to FIG. 15A.

A serial correlator 1504 may be designed to operate with multi-bitresolution to improve correlation accuracy.

FIG. 22 is a block diagram showing a preferred embodiment of a Robertsondevice 1601.

The Robertson device 1601 has an input 1602 and an input 1603, andcomputes an approximation of the square root of the sum of the squaresof its inputs, as shown in equation 1152. The input 1602 and input 1603may be binary inputs such as 5-bit binary numbers. The input 1602 andthe input 1603 are coupled to a comparator 1604, which generates acontrol output 1605 indicating whether the input 1602 is greater thanthe input 1603.

The input 1602 and the input 1603 are also coupled to a selector 1606,which outputs the greater of the input 1602 and the input 1603 inresponse to the control output 1605.

The input 1602 and the input 1603 are also coupled to a selector 1607,which outputs the lesser of the input 1602 and the input 1603 inresponse to an inverse of the control output 1605.

The output of the selector 1606 and the output of the selector 1607 arecoupled to an adder 1608. However, prior to being connected to the adder1608, the output of the second selector 1607 is shifted right one bit,i.e., the 0 (least significant) bit of the output of the second selector1607 may be discarded, the 1 (next least significant) bit of the outputof the second selector 1607 may be transferred to the 0 (leastsignificant) bit position, the 2 bit of the output of the secondselector 1607 may be transferred to the 1 bit position, and so on. Theright shift has the effect of dividing the output of the second selector1607 by two (and dropping the least significant bit).

The output of the adder 1608 may be output from the Robertson device1601, which therefore effectuates equation 1152 as set forth herein.

Previously herein is explained the concept and operation of M-ary spreadspectrum transmission, whereby data throughput may be increased byassigning a different predefined data bit pattern to each of M differentspreading codes (i.e., symbol codes) and deriving the predefined databit pattern at the receiver in response to determining which of the Msymbol codes was transmitted. Thus, for example, the receiverembodiments shown in FIGS. 18, 19, 21A and 21B have been describedpreviously with reference to a 32-ary system, wherein 32 correlatorsoperate in parallel to determine which of 32 symbol codes has beentransmitted, and to derive one of the predefined data symbols thereby.Data throughput may be further increased by use of phase encoding asdescribed below.

Phase encoding generally involves the imposition in the transmittedsignal of known phase changes at selected intervals, wherein the phasechanges correspond to information to be transmitted apart from or inaddition to the M-ary encoded information. Decoding of the phase changesat the receiver allows recognition of the phase encoded information.

Phase encoding may be either absolute or differential in nature.Absolute phase encoding generally involves the imposition of a selectedphase upon the signal to be transmitted irrespective of the immediatelyprior phase of the transmitted signal. Differential phase encodinggenerally involves the imposition of a selected phase upon the signal tobe transmitted, giving consideration to the immediately prior phase ofthe transmitted signal. For absolute phase encoding, recovery andtracking of the carrier signal is usually necessary at the receiver,which may involve a difficult and relatively complex process. To avoidhaving to recover and track the carrier signal, differential phaseencoding is generally preferred over absolute phase encoding withrespect to the embodiments disclosed herein.

FIGS. 24A and 24B are digital circuit block diagrams of a spreadspectrum transmitter employing differential phase encoding, and FIG. 24Cis an abstract block diagram of the transmitter of FIGS. 24A and 24B. InFIG. 24C, a data signal 2461 comprising a plurality of data bits isserially clocked into registers 2462 and 2463. The data bits in register2462 form a data symbol, such as previously described herein withrespect to the transmitter of FIG. 17A, and comprise an address 2464 foraccessing a symbol table 2466. Symbol table 2466 comprises a pluralityof spread spectrum codes, or symbol codes, such as also previouslydescribed herein with respect to the transmitter of FIG. 17A. Inresponse to each data symbol in register 2462, a symbol code from symboltable 2466 is selected and output over line 2475.

The data in register 2463 comprises phase encoding information. In apreferred embodiment, register 2463 comprises a single-bit register orflip-flop, and therefore holds one data bit of information from datasignal 2461.

Register 2463 is connected to one input of an XOR gate 2472. A priorphase state register 2470 holds the prior phase state information,θ_(j−1), and is connected to the other input of XOR gate 2472. In oneembodiment, prior phase state register 2470 holds a 0-bit value if theprevious phase was 0°, and holds a 1-bit value if the previous phase was180°. The present phase state θ_(j) is selected based on the prior phasestate information θ_(j−1) stored in prior phase state register 2470 andthe phase encoding bit stored in register 2463, according to a preferredencoding scheme shown in Table 24-1.

TABLE 24-1 Previous Code Previous Phase Encoding Present Code Phase --θ_(j−1) Representation Bit Phase -- θ_(j)  0° 0 0  0°  0° 0 1 180° 180°1 0 180° 180° 1 1  0°

where the previous phase representation is stored in register 2470, andthe encoding bit is stored in register 2463. The phase of thetransmitted signal remains the same if register 2463 contains a 0-bitvalue, but is inverted (i.e., by taking the complement of each chip inthe symbol code) if register 2463 contains a 1-bit value. The XOR gate2472 thereby selects a present phase state θ_(j), and outputs a phaseselection signal 2477 according to the logic shown in Table 24-1. Aftereach symbol code period, the present phase state θ_(j) from phaseselection signal 2477 is stored in the prior phase state register 2470.The phase selection signal 2477 is coupled to a phase selector 2476. Thephase selector 2476 operates on the symbol code selected from symboltable 2466 according to the logic of Table 24-1. Thus, phase selector2476 inverts the selected symbol code if the output of XOR gate 2472 isa 1, and does not invert the selected symbol code if the output of XORgate 2472 is a 0. The phase selector 2476 outputs a phase encoded signal2479. The phase encoded signal 2479 may be sent to a modulator forfurther processing, such as for dividing into I and Q chip streams,generating I and Q waveforms in response to the I and Q chip streams,and combining and transmitting the I and Q waveforms in a manner similarto that described generally with respect to the transmitter of FIG. 6.

In an exemplary embodiment, the transmitter of FIG. 24C operates in a32-ary system, wherein each of 32 spread spectrum codes or symbol codeseach represent a different data symbol, and each data symbol comprises aunique pattern of 5 data bits. Six bits are sent in each symbol period.Five bits are used to select a symbol code, while the sixth bit is usedto differentially encode the symbol code. In this exemplary embodiment,40 symbols are sent per transmission burst in a time division multipleaccess communication system such as described previously, each symbol(except the first symbol) conveying six bits of information, includingthe phase encoded information. Thus, a total of 239 bits of informationare transmitted per transmission burst, an almost 20% increase in datathroughput over non-phase encoded transmission.

The first symbol code in each transmission burst acts as a phasereference, and therefore conveys no phase encoded information. Thus, inthe exemplary embodiment described above, the first symbol code conveysonly five bits of information. Subsequent symbol codes are phase encodedand therefor convey six bits of information for each symbol code.

FIG. 24D is a diagram of an exemplary input data sequence and anexemplary symbol code output sequence. In FIG. 24D, a data sequence 2490comprising data bits 2491 corresponds, for example, to data signal 2461in FIG. 24C. A specific exemplary data sequence 2492 with actual datavalues is also shown in FIG. 24D, with relation to data sequence 2490. Afirst symbol S1 corresponds to the first five bits B0-B4 in the datasequence 2492; a second symbol S2 corresponds to the next five bitsB5-B9 in the data sequence 2492; and so on. The phase of the firstsymbol S1 establishes a reference. The phase reference may be selectedas, e.g., 0°. The phase of the second symbol S2 is determined by thesixth bit (i.e., bit B10) after the first symbol S1, according to thelogic set out in Table 24-1. Because in the present example bit B10 is a1-bit, the phase of the second symbol S2 is inverted with respect tofirst symbol S1—i.e., the phase of the second symbol S2 is 180°.

In a similar manner, for a third symbol S3 corresponding to the nextfive bits B11-B15 in the data sequence 2492, and its phase isestablished by the sixth bit B16 following the preceding data symbol S2.Because in the present example bit B16 is a 0-bit, the phase of thethird symbol S3 is not inverted with respect to the second symbolS2—i.e., the phase of third symbol is also 180°. The same encodingselection is performed for the subsequent bits in the data sequence2492, with each five bits of a six-bit sequence 2494 defining a symbol,and the sixth bit 2493 of the six-bit sequence 2494 defining therelative phase of the symbol.

An output signal 2497 in FIG. 24D comprises a sequence of phase-encodeddata symbol codes 2495. Thus, for the exemplary data sequence 2492, theoutput signal 2497 comprises a non-inverted fifth symbol code M5, aninverted seventeenth symbol code M17, an inverted twenty-fourth symbolcode M24, a non-inverted fourth symbol code M4, and so on.

FIGS. 24A and 24B are more detailed digital circuit block diagrams of aspread spectrum transmitter employing differential phase encoding. Inthe embodiment shown in FIGS. 24A and 24B, a serial input data stream2401 is coupled to a CRC (cyclical redundancy check) encoder 2402. TheCRC encoder 2402 adds bits to the serial input data stream 2401 whichmay be used at a receiver for determining if a transmitted signal wassent without errors. The CRC encoder 2402 outputs a serial data signal2403, which corresponds, e.g., to data signal 2461 in FIG. 24C.

Data signal 2403 is coupled to a serial-to-parallel register 2404, whichconverts the data signal 2403 into a series of 6-bit sequences. Thefirst five bits of each 6-bit sequence are connected over lines 2405 toa latch 2407. The sixth bit of each 6-bit sequence, referred to hereinas the phase selection bit, is coupled over line 2406 to a symbol phaseencoder 2413. Output lines 2408 from latch 2407 are used to select oneof M symbols stored in a symbol code lookup table 2444 (e.g., a ROM)shown in FIG. 24B. In an exemplary embodiment, the symbol code lookuptable 2444 stores the set of thirty-two symbol codes appearing in Table17-4. Output lines 2408 comprise five bits of an address of the symbolcode lookup table 2444. The lookup table address also comprises chipcount lines 2441 received from a chip counter 2440.

In operation, data bits in data signal 2403 are clocked into theserial-to-parallel register 2404 under control of a clock signal 2425(e.g., a 5 MHz clock). The contents of the serial-to-parallel register2404 are loaded in parallel to latch 2407 once each symbol period. Aload latch signal 2409 controls loading of the latch 2407, such that thelatch 2407 is loaded when both a transmit enable signal 2460 and an endsymbol signal 2453 are active. The transmit enable signal 2460 isactivated by a processor or other controller (not shown) when it isdesired to transmit data over a communication channel. The end symbolsignal 2453 is generated by an AND gate 2452 (shown in FIG. 24B) whichreceives as inputs the chip count lines 2441 and produces an activeoutput when all of the chip count lines 2441 are in a logical highstate—that is, chip counter 2440 has finished counting up to 32.

As noted, the output lines 2408 of the latch 2407 and chip count lines2441 are used as an address for the symbol code lookup table 2444.Preferably, lines 2408 comprise the most significant bits of theaddress, and chip count lines 2441 comprise the least significant bitsof the address. Each clock period of the clock signal 2415, the clockcounter 2440 increments its count, thereby cycling through thirty-twodifferent states reflected in the binary count on chip count lines 2441.In response to the ten address lines (five symbol selection lines 2408and five chip count lines 2441), the symbol code lookup table 2444outputs a symbol code signal 2446 comprising a sequence of chipscorresponding to the selected symbol code. Each time the clock counter2440 increments, chip count lines 2441 change accordingly and accessesthe next chip of the selected symbol code stored in the symbol codelookup table 2444.

Differential phase encoding of the symbol code signal 2446 isaccomplished by performing an exclusive-OR operation with a phaseselection signal 2418 output from the phase encoder 2413 and the symbolcode signal 2446 using XOR gate 2447 in FIG. 24B. The phase encoder 2413operates by taking the phase selection bit of each 6-bit sequence fromline 2406, as noted above, and comparing it with the previous phase asstored in a previous phase register 2412 (e.g., a flip-flop). In phaseencoder 2413, XOR gate 2410 and previous phase register 2412 correspondfunctionally with XOR gate 2472 and flip-flop 2470 in FIG. 24C, exceptthat the order of XOR gate 2410 and previous phase register 2412 arereversed to preserve synchronous operation. Loading of previous phaseregister 2412 is controlled by end symbol signal 2453. At the same timelatch 2407 is loaded with the new data symbol, previous phase register2412 is loaded with the new phase. While the symbol code is beingtransmitted, the next data symbol may be loaded into serial-to-parallelregister 2404, and the next phase determined by XOR gate 2410. At theend of the symbol code transmission, the next data symbol and the nextphase are loaded into latch 2407 and previous phase register 2412,respectively.

The output of previous phase register 2412 is a phase state signal 2414.Phase state signal 2414 is gated with a phase enable signal 2415. Whenthe phase enable signal 2415 is active, symbol code signal 2446 isdifferentially phase encoded, and the transmitter thereby sends six bitseach symbol period; when the phase enable signal 2415 is inactive,symbol code signal 2446 is not phase encoded, and the transmitterthereby sends only five bits each symbol period.

The output of XOR gate 2447, which outputs a differentially phaseencoded symbol code signal 2461 when phase enable signal 2415 is active,is connected to a multiplexer 2449. In response to a select signal 2448,multiplexer 2449 selects as an output either the differentially phaseencoded symbol code signal 2461, or a preamble/fill code signal 2462from a preamble/fill code table 2443. The preamble/fill code table 2443stores a preamble code comprising, e.g., 48 chips, and a fill codecomprising, e.g., 16 chips, for a total of 64 chips. The preamble/fillcode table 2443 is addressed by chip count lines 2441 and a sixth line2463, to allow sixty-four stored chips to be sequentially accessed.

In a preferred embodiment, for a given burst in accordance with the TDMAtiming structure shown, e.g., in FIG. 17D, select signal 2448 firstselects as an output sixty-four chips comprising a preamble and fillcode from preamble/fill code table 2443. After sixty-four chips areoutput, the select signal 2448 changes state and selects as an outputthe differentially encoded symbol code signal 2461. In a particularembodiment, the select signal 2448 selects forty symbols to betransmitted from the differentially encoded symbol code signal 2461.Multiplexer 2450 outputs a chip stream signal 2461 to a modulator,wherein the chip stream signal 2461 may be divided into I and Q chipstreams for generating and transmitting of a CPM signal as previouslydescribed herein.

FIGS. 25A and 25B-25C are block diagrams of two different embodiments ofa receiver for recognizing phase information in a receiveddifferentially phase encoded CPM signal. In FIG. 25A, a receiver 2501comprises a CPM correlator 2502 which generates a real correlationsignal 2511 and an imaginary correlation signal 2512 in response toreceiving a phase encoded CPM signal. The CPM correlator 2502 of FIG.25A may be embodied as any of the CPM correlators of FIGS. 10, 12, 14,15A, 15B or 15D which generate real and imaginary correlation signals.In the particular embodiment shown FIG. 25A, the correlator of FIG. 15Ais used.

The real correlation signal 2511 and the imaginary correlation signal2512 are coupled to a phase discriminator 2510 which, in responsethereto, determines the phase angle of the received signal. In apreferred embodiment, the phase discriminator 2510 determines not theprecise phase angle of the received signal, but only a sector which thephase angle lies within. Operation of the phase discriminator 2510 maybe explained with reference to FIG. 27A. FIG. 27A is a phase angle graphshowing a circle 2701 divided into a plurality of sectors 2702. Thex-axis of the graph of FIG. 27A corresponds to a real correlation value,while the y-axis of the FIG. 27A graph corresponds to an imaginarycorrelation value. Assuming a lossless communication channel and acapability of perfect correlation, the real correlation value andimaginary correlation value may be seen as forming coordinates <Re, Im>for each symbol that would lie somewhere on circle 2701. In other words,the total correlation magnitude C for a correlated symbol would alwaysbe the same (Re²+Im²=C²), but the phase angle would vary along thecircle 2701 depending on the relative phase difference in thetransmitter and receiver clocks.

Because it may be assumed that the communication channel will be subjectto losses and noise interference, and that the correlator hardware haspractical limitations, the total correlation magnitude C for acorrelated symbol may be other than the total correlation valuerepresented by circle 2701. Thus, the real correlation value andimaginary correlation value coordinates <Re, Im> may generally lieanywhere within or even without the circle 2701.

The phase discriminator 2510 determines the phase of the received CPMsignal by determining the sign of the real correlation signal 2511 andthe sign of the imaginary correlation signal 2512, and by comparing therelative magnitudes of the real correlation signal 2511 and theimaginary correlation signal 2512. Based on the derived information, thephase discriminator 2510 determines the sector 2702 in which the phaseangle lies.

In more detail, the real correlation signal 2511 is compared againstzero by comparator 2517, which outputs a real sign signal 2523. Theimaginary correlation signal 2512 is compared against zero by comparator2515, which outputs an imaginary sign signal 2521. The relativemagnitudes of the real correlation signal 2511 and the imaginarycorrelation signal 2512 are compared by a magnitude comparator 2516,which outputs a magnitude comparison signal 2522. The magnitudecomparator 2516 and comparators 2515 and 2517 may be either analog ordigital, depending upon whether the real correlation signal 2511 andimaginary correlation signal 2512 are analog or digital signals.

The real sign signal 2523, imaginary sign signal 2521, and magnitudecomparison signal 2522 are connected to a sector logic block 2530, whichoutputs a phase sector signal 2531 identifying the sector 2702 of thereceived phase angle as shown in FIG. 27A. The sectors 2702 in FIG. 27Aare arranged as follows. Each sector 2702 covers a 45° region of circle2701, with each pair of adjacent sectors 2702 defining a quadrant. Thus,sectors 0 and 1 define a first quadrant; sectors 2 and 3 define a secondquadrant; sectors 4 and 5 define a third quadrant; and sectors 6 and 7define a fourth quadrant. The real sign signal 2523 and imaginary signsignal 2521 together determine the quadrant of the phase angle, whilethe magnitude comparison signal 2522 determines which sector 2702 of thequadrant the phase angle lies in.

Thus, for example, where the sign of the real correlation signal 2511and the sign of the imaginary correlation signal 2512 are both positive,it may be concluded that the phase angle lies in the quadrant defined bysectors 0 and 1. The magnitude comparison signal 2522 then determineswhich of sector 0 and 1 the phase angle lies within. If the realcorrelation signal 2511 (i.e., the first coordinate Re of the <Re, Im>pair) is equal in magnitude to the imaginary correlation signal 2512(i.e., the second coordinate Im of the <Re, Im> pair), then the phaseangle would lie on the 45° border between sectors 0 and 1. If the realcorrelation signal 2511 is greater in magnitude than the imaginarycorrelation signal 2512, then the phase angle lies below the 4520 borderbetween sectors 0 and 1 and therefore lies in sector 0. Similarly, ifthe real correlation signal 2511 is smaller in magnitude than theimaginary correlation signal 2512, then the phase angle lies above the45° border between sectors 0 and 1 and therefore lies in sector 1.

Table 25-1 illustrates the eight possible combinations of realcorrelation signal sign, imaginary correlation signal sign, and relativemagnitude of real and imaginary correlation signals for the sectorarrangement of FIG. 27A.

TABLE 25-1 Larger Real Sign Imaginary Sign Magnitude Sector − − Re 4 − −Im 5 − + Re 3 − + Im 2 + − Re 7 + − Im 6 + + Re 0 + + Im 1

Phase logic block 2530 implements Table 25-1, and, in response to itsinputs, outputs a three-bit phase sector signal 2531 identifying thesector in which the phase angle lies.

Once the sector of the phase angle is determined, the phase informationof the received signal may be decoded by comparing the current phasesector against the previous phase sector. If the current phase sectordiffers from the previous phase sector by an amount closer to 0° than180°, then it may be concluded that there was no phase change in thereceived signal and, therefore, that the phase information encoded inthe received signal is a 0-bit. Conversely, if the current phase sectordiffers from the previous phase sector by an amount closer to 180° than0°, then it may be concluded that there was a phase inversion in thereceived signal and, therefore, that the phase information encoded inthe received signal is a 1-bit.

The phase sector comparison may be further explained with reference toFIG. 27A. As an example, assume that the previous phase sector wassector 0. In such a case, if the current phase sector is any of sectors0, 1 or 7, then it may be concluded that there was no phase change inthe received signal and, therefore, that the phase information encodedin the received signal is a 0-bit. If, on the other hand, the currentphase sector is any of sectors 3, 4 or 5, then it may be concluded thatthere was a phase inversion in the received signal and, therefore, thatthe phase information encoded in the received signal is a 1-bit. If,however, the current phase sector is either sector 2 or 6, it cannotnecessarily be concluded with sufficient certainty whether or not aphase inversion occurred in the received signal. The reason for thisambiguity is that the phase angle is approximated each symbol period interms of a 45° sector, and is not measured to a finer degree. Experimenthas shown that if the current phase sector falls in either of thesectors at a 90° orientation with respect to the previous phase sector,then treating the situation as one in which there is no phase inversionis preferred. Thus, in the present example, if the current phase sectoris either sector 2 or 6, then the phase change should be treated as 0°,and the phase information considered a 0-bit.

More generally, if the current phase sector is positioned within twosectors 2702 of the previous phase sector, then it may be concluded thatno phase change has occurred in the received signal. If, on the otherhand, the current phase sector is positioned more than two sectors 2702away from the previous phase sector, then it may be concluded that aphase inversion has occurred in the received signal.

FIGS. 25B and 25C are block diagrams of an alternative embodiment of areceiver having a phase decoding capability such that phase informationin a received differentially phase encoded CPM signal may be recognized.In FIG. 25B, a receiver 2551 comprises a CPM correlator 2552 whichgenerates a real correlation signal 2561 and an imaginary correlationsignal 2562 in response to receiving a phase encoded CPM signal. The CPMcorrelator 2552 of FIG. 25B may be embodied as any of the CPMcorrelators of FIGS. 10, 12, 14, 15A, 15B or 15D which generate real andimaginary correlation signals. In the particular embodiment shown FIG.25B, the correlator of FIG. 15A is used.

The real correlation signal 2561 and the imaginary correlation signal2562 are coupled to a phase discriminator 2560 which, in responsethereto, determines the phase angle of the received signal. In apreferred embodiment, the phase discriminator 2560 determines not theprecise phase angle of the received signal, but only a sector which thephase angle lies within. Operation of the phase discriminator 2560 maybe explained with reference to FIG. 27B. FIG. 27B is a phase map showinga circle 2721 divided into a plurality of sectors 2722, similar to FIG.27A. Phase discriminator 2560 determines which sector 2722 the phaseangle of the received signal lies in, and is therefore functionallysimilar to phase discriminator 2510 of FIG. 25A.

In a preferred embodiment, the real correlation signal 2561 andimaginary correlation signal 2562 are derived using integrators 2553 and2554, respectively, wherein integrators 2553 and 2554 each comprise adigital counter. Thus, integrators 2553 and 2554 each output a binarycount signal representing a correlation value, such as a 5-bit binarysignal. Real correlation signal 2561 and imaginary correlation signal2562 are each connected to a truncate block 2565, which preferablyselects a predefined number of the most significant bits of its inputs.

In a particular embodiment, integrators 2553 and 2554 each comprisedigital up-counters, and real correlation signal 2561 and imaginarycorrelation signal 2562 each comprise a first sign bit followed by fourmagnitude bits. In this embodiment, a correlation value of thirty-one(binary 11111) represents a maximum positive correlation, a correlationvalue of fifteen (binary 01111) or sixteen (binary 10000) represents aminimal correlation, and a correlation value of zero (binary 00000)represents a maximum negative correlation. In a preferred embodiment,integrators 2553 and 2554 are embodied as six-bit digital counters so asto reach a maximum positive correlation value of thirty-two (binary100000) instead of thirty-one.

In the FIG. 25B embodiment, truncate block 2565 selects the three mostsignificant bits of the real correlation signal 2561 and the three mostsignificant bits of the imaginary is correlation signal 2562. The phasediscriminator 2560 uses these truncated correlation values to estimatethe phase angle according to the general equation φ=Arctan(Im/Re).Because each truncated correlation value represents a range ofcorrelation values, a median value is chosen for each truncated valuefor use in the Arctangent calculation. In a preferred embodiment, themedian value chosen for each truncated value is selected according toTable 25-2.

TABLE 25-2 VALUE CORRELATION USED FOR RANGE OVER SCORE ARCTAN WHICHTRUNCATED (SIGN, 2 MSB's) CALCULATION SCORE CAN VARY 000 −14 −12→−15 001−10  −8→−11 010 −6 −4→−7 011 −2  0→−3 100 2 0→3 101 6 4→7 110 10  8→11111 14 12→15

By using three bits from the real correlation signal 2561 and three bitsfrom the imaginary correlation signal 2562 to estimate the phase angle,the phase angle is thereby quantized into one of sixty-four possiblelocations in the phase map of FIG. 27B. The different possible phaseangles and resulting sector location may be determined according toTable 25-3 below, wherein “Real”represents the truncated realcorrelation value, “Imag”represents the truncated imaginary correlationvalue, “Real Vector Value” is the median real correlation value selectedbased on the truncated real correlation value according to Table 25-2,“Imag Vector Value” is the median imaginary correlation selected basedon the truncated imaginary correlation value according to Table 25-2,“Phase” is the phase angle calculated based on an arctangent of the RealVector Value and the Imag Vector Value, and “Sector” refers to thesector in which the phase angle lies, according to a preferred sectormapping shown in FIG. 27C.

TABLE 25-3 SECTOR MAPPING REAL IMAG VECTOR VECTOR VALUE REAL VALUE IMAGSECTOR PHASE −14 000 −14 000 A 225 −14 000 −10 001 A 215 −14 000 −6 0109 200 −14 000 −2 011 8 188 −14 000 +2 100 8 172 −14 000 +6 101 7 160 −14000 +10 110 6 145 −14 000 +14 111 6 135 −10 001 −14 000 A 234 −10 001−10 001 A 225 −10 001 −6 010 9 210 −10 001 −2 011 9 191 −10 001 +2 100 7169 −10 001 +6 101 7 150 −10 001 +10 110 6 135 −10 001 +14 111 6 125  −6010 −14 000 B 247  −6 010 −10 001 B 239  −6 010 −6 010 A 225  −6 010 −2011 9 198  −6 010 +2 100 7 162  −6 010 +6 101 6 135  −6 010 +10 110 5121  −6 010 +14 111 5 113  −2 011 −14 000 C 262  −2 011 −10 001 C 259 −2 011 −6 010 B 251  −2 011 −2 011 A 225  −2 011 +2 100 6 135  −2 011+6 101 5 109  −2 011 +10 110 4 101  −2 011 +14 111 4  98  +2 100 −14 000C 278  +2 100 −10 001 C 281  +2 100 −6 010 D 288  +2 100 −2 011 E 315 +2 100 +2 100 2  45  +2 100 +6 101 3  72  +2 100 +10 110 4  79  +2 100+14 111 4  82  +6 101 −14 000 D 293  +6 101 −10 001 D 301  +6 101 −6 010E 315  +6 101 −2 011 F 342  +6 101 +2 100 1  18  +6 101 +6 101 2  45  +6101 +10 110 3  59  +6 101 +14 111 3  67 +10 110 −14 000 E 305 +10 110−10 001 E 315 +10 110 −6 010 F 329 +10 110 −2 011 F 349 +10 110 +2 100 1 11 +10 110 +6 101 1  31 +10 110 +10 110 2  45 +10 110 +14 111 2  55 +14111 −14 000 E 315 +14 111 −10 001 E 325 +14 111 −6 010 F 340 +14 111 −2011 0 352 +14 111 +2 100 0  8 +14 111 +6 101 1  20 +14 111 +10 110 2  35+14 111 +14 111 2  45

FIG. 27C is a diagram of a preferred sector mapping. FIG. 27C shows acircle 2741 (similar to the circle 2721 of FIG. 27B) comprising aplurality of sectors 2742. Circle 2741 is divided into sectors 2742denoted sector 0, 1, 2, . . . F, according to the mapping set forth inTable 25-4 below.

TABLE 25-4 SECTOR DEFINITION PHASE SECTOR 349-11  0 11-33 1 33-56 256-78 3  78-101 4 101-124 5 124-146 6 146-168 7 168-191 8 191-214 9214-236 A 236-259 B 259-281 C 281-304 D 304-326 E 326-349 F

In a preferred embodiment, the sector for the current phase angle isdetermined by using a six bit signal 2566 comprising the truncated realcorrelation signal and the truncated imaginary correlation signal as anaddress 2570 for a sector lookup table 2571. The sector lookup table2571 may comprise, e.g., a ROM or other non-volatile memory, and outputsa four-bit binary sector signal 2573 indicating which of the sixteensectors 2742 the phase angle lies in. In a preferred embodiment, thecontents of the sector lookup table 2571 are selected according to Table25-5.

TABLE 25-5 SECTOR ROM CONTENTS ADDRESS (hex) DATA (hex) (Re, Im)(Sector) 00 A 01 A 02 9 03 8 04 8 05 7 06 6 07 6 08 A 09 A 0A 9 0B 9 0C7 0D 7 0E 6 0F 6 10 B 11 B 12 A 13 9 14 7 15 6 16 5 17 5 18 C 19 C 1A B1B A 1C 6 1D 5 1E 4 1F 4 20 C 21 C 22 D 23 E 24 2 25 3 26 4 27 4 28 D 29D 2A E 2B F 2C 1 2D 2 2E 3 2F 3 30 E 31 E 32 F 33 F 34 1 35 1 36 2 37 238 E 39 E 3A F 3B 0 3C 0 3D 1 3E 2 3F 2

Once the current sector is determined, the phase information from thereceived signal may be recognized in a manner similar to that describedwith respect to FIG. 25A. A preferred embodiment of phase decodingcircuitry is shown in FIG. 25C. In FIG. 25B, and in more detail in FIG.25C, are shown address lines 2570 connected to a sector lookup table2571, which outputs a sector signal 2573. FIG. 25C further shows sectorsignal 2573 connected to a register 2580, which stores the previoussector value. A previous sector signal 2581 is output from register 2580and connected to one set of inputs of a subtractor 2585, and sectorsignal 2573 is connected to another set of inputs of the subtractor2585. Subtractor 2585 subtracts its inputs and generates a sectordifference signal 2586.

The sector difference signal 2586 is used to derive the encoded phaseinformation. If the current phase sector is positioned within foursectors 2742 of the previous phase sector, then it may be concluded thatno phase change has occurred in the received signal and, therefore, thatthe phase information encoded in the received signal is a 0-bit. If, onthe other hand, the current phase sector is positioned more than foursectors 2742 away from the previous phase sector, then it may beconcluded that a phase inversion has occurred in the received signaland, therefore, that the phase information encoded in the receivedsignal is a 1-bit. Accordingly, the sector difference signal 2586 isapplied as an address to a phase bit lookup table 2590, which outputs aphase bit signal 2591 comprising a 0-bit or a 1-bit depending on thevalue of the sector difference signal 2586. In a preferred embodiment,the phase bit lookup table 2590 comprises a ROM or other non-volatilememory, the contents of which are in accordance with Table 25-6.

TABLE 25-6 PHASE ROM CONTENTS ADDRESS (hex) DATA (hex) (SectorDifference) (Nth Bit) 0 0 1 0 2 0 3 0 4 0 5 1 6 1 7 1 8 1 9 1 A 1 B 1 C0 D 0 E 0 F 0

It may be noted that the 16-sector embodiment of FIG. 27C, like the8-sector embodiment of FIG. 27A, has two sectors 2742 of ambiguity whichare aligned at 90° to the previous phase sector. But because there aremore sectors 2742 in the FIG. 27C embodiment than in the FIG. 27Aembodiment, and hence a narrower sector size, the regions of ambiguityare reduced in the FIG. 27C embodiment. By increasing the number ofsectors (which may be done, e.g., by increasing the number of bits usedfrom the correlation signals 2561 and 2562 to calculate the phaseangle), the sector size can be further narrowed, so as to further reducethe total region of ambiguity. As with the FIG. 27A embodiment, a phasedifference falling in a region of ambiguity is preferably treated asindicating no phase inversion—i.e., the phase information is treated asa 0-bit.

FIG. 26 is a block diagram of a preferred receiver for carrying outphase decoding in a 32 symbol transmission technique in accordance withthe embodiment of the receiver shown in FIGS. 25B and 25C. In FIG. 26, areceived signal 2605 is coupled to a plurality of CPM correlators 2610(e.g., 32 different correlators). Each of the CPM correlators 2610 maybe embodied as any of the CPM correlators of FIGS. 10, 12, 14, 15A, 15Bor 15D, and each CPM correlator 2610 simultaneously outputs a realcorrelation signal 2612, an imaginary correlation signal 2613, and aunified correlation signal 2611 in response to receiving the incomingsignal 2605. In a preferred embodiment, each of correlators 2610comprises a correlator such as shown in FIG. 15D.

The correlation signal 2611 from each of the CPM correlators 2610 iscoupled to a best-of-M detector 2620, which compares the relativemagnitudes of each of the unified correlation signals 2611 and selectsthe one indicating the highest degree of correlation. The best-of-Mdetector 2620 outputs a signal 2621 indicating which of the thirty-twosymbols has the highest degree of correlation. Signal 2621 is coupled asa select control signal to a real correlation signal multiplexer 2625and an imaginary correlation signal multiplexer 2626. The realcorrelation signals 2612 from each of the CPM correlators 2610 areconnected as inputs to the real correlation signal multiplexer 2625, andthe imaginary correlation signals 2613 from each of the CPM correlators2610 are connected as inputs to the imaginary correlation signalmultiplexer 2626. In response to signal 2621, the real correlationsignal 2612 and the imaginary correlation signal 2613 corresponding tothe highest correlation symbol are output from the real correlationsignal multiplexer 2625 and the imaginary correlation signal multiplexer2626, respectively, as a selected real correlation signal 2627 and aselected imaginary correlation signal 2628.

The selected real correlation signal 2627 and the selected imaginarycorrelation signal 2628 are connected to a phase computation block 2630.The phase computation block 2630 outputs a phase estimate signal 2631which is connected to a previous phase estimate memory 2635 and asubtractor 2640. The subtractor 2640 calculates a difference between thephase estimate signal 2631 and a previous phase estimate signal 2636stored in the previous phase estimate memory 2635, and derives a phasedifference signal 2641 thereby. The phase difference signal 2641 isconnected to a magnitude comparator 2642 which determines in responsethereto the phase encoded information. The phase computation block 2630,previous phase estimate memory 2635, subtractor 2640, and magnitudecomparator 2624 may generally be embodied as sector lookup table 2571,register 2580, subtractor 2585, and phase bit lookup table 2590appearing in FIG. 25C.

The techniques described above with respect to single bit or biphaseencoding may be applied to other levels of encoding, such as, e.g.,triphase, quadraphase or octiphase encoding. In quadraphase encoding,for example, two bits of the data signal in the transmitter are used forphase encoding. For each symbol, the phase may be in any one of fourrelative states, each at 90° with respect to the previous phase state.The phase angle may be determined previously described with respect toFIGS. 25A-25C. Depending upon the relative phase difference as reflectedin the current and previous sector values, one of four phase states maybe derived, and two bits of phase information data recovered in responseto the selected one of four phase states.

Alternative Embodiments

While preferred embodiments are disclosed herein, many variations arepossible which remain within the concept and scope of the invention, andthese variations would become clear to one of ordinary skill in the artafter perusal of the specification, drawings and claims herein.

In one alternative embodiment, the circuitry constituting either FIG.17, or FIGS. 18, 19, and 21A-B, or all said figures, may be incorporatedonto a single integrated chip, along with supporting circuitry asnecessary. Also, while information to be transmitted from transmitter toreceiver is generally referred to herein as “data”, the term “data” maycomprise data, error-correcting codes, control information, protocolinformation, or other signals, and all these are deemed to be within thescope and spirit of the invention.

While the invention as shown in embodiments herein uses certain CPMencoding techniques, those skilled in the art would recognize, afterperusal of this application, that a number of encoding methods, such asMSK, GMSK, SQAM, SQORC, and other known spread-spectrum techniques,would be workable and fall within the scope and spirit of the invention.The invention therefore is not to be restricted except within the spiritand scope of the appended claims.

We claim:
 1. A method for spread-spectrum communication between a firsttransceiver and a second transceiver, the method comprising the stepsof: transmitting, from a spread spectrum transmitter associated with thefirst transceiver to a spread spectrum receiver associated with thesecond transceiver, a spread spectrum preamble; transmitting, from saidspread spectrum transmitter to said spread spectrum receiver, a spreadspectrum encoded data message immediately following said spread spectrumpreamble, said spread spectrum encoded data message separated in timefrom said spread spectrum preamble by a predefined gap during which thesecond transceiver does not transmit to the first transceiver; receivingsaid spread spectrum preamble at said spread receiver; generating, atsaid spread spectrum receiver, a correlation signal in response to saidspread spectrum preamble; receiving said spectrum encoded data messageat said spread spectrum receiver; and despreading said spread spectrumencoded data message at said spread spectrum receiver.
 2. The method ofclaim 1, wherein said predefined gap is of sufficient duration to allowcorrelation of said spread spectrum preamble by said spread spectrumreceiver.
 3. The method of claim 1, further comprising the steps ofgenerating a repeating time frame, and defining a plurality of equallyspaced time offsets relative to a starting boundary of said repeatingtime frame, wherein said step of transmitting said spread spectrumpreamble comprises the step of transmitting said spread spectrumpreamble with reference to one of said time offsets.
 4. A spreadspectrum communication system, comprising: a first transceiver having afirst spread spectrum transmitter and a first spread spectrum receiver,said first spread spectrum transmitter transmitting a burst comprising aspread spectrum preamble and a spread spectrum encoded data messagefollowing said spread spectrum preamble after a preamble sounding gap;and a second transceiver having a second spread spectrum transmitter anda second spread spectrum receiver, said second spread spectrum receivercomprising circuitry for receiving and correlating to said spreadspectrum preamble and circuitry for receiving and processing said spreadspectrum encoded data message; wherein said second spread spectrumtransmitter does not transmit to said first spread spectrum receiverduring said preamble sounding gap.
 5. A spread spectrum communicationsystem, comprising: a first transceiver comprising a spread spectrumtransmitter; and a second transceiver comprising a spread spectrumreceiver; wherein said spread spectrum transmitter and said spreadspectrum receiver communicate using an over-the-air protocol accordingto which said spread spectrum transmitter transmits, and said spreadspectrum receiver receives, a spread spectrum preamble followed after anidle period of predetermined duration by a spread spectrum encoded datamessage, said idle period being relatively short in comparison to aperiod of transmitting said spread spectrum encoded data message, noinformation being transmitted by said spread spectrum transmitter orreceived by said first transceiver during said idle period.
 6. Thespread spectrum communication system of claim 5, wherein said spreadspectrum transmitter transmits a non-informational fill code during saididle period.
 7. A multiple-access spread spectrum communication system,comprising: a base station, said base station comprising a base stationtransmitter and a base station receiver; and a plurality of userstations, each of said user stations comprising a user stationtransmitter and a user station receiver; wherein each of said userstations transmits on an assigned channel a burst with reference to oneof a plurality of time slots of a repeating time frame during whichburst the user station does not receive information from the basestation on the same channel, each burst comprising a spread spectrumpreamble, a preamble sounding gap following said spread spectrumpreamble, and a spread spectrum encoded data message following saidspread spectrum preamble, said preamble sounding gap being shorter induration than said spread spectrum encoded data message; and wherein,for each burst, said base station receives said spread spectrumpreamble, correlates said spread spectrum preamble during said preamblesounding gap, and despreads said spread spectrum encoded data messagethereafter based upon the correlation of said spread spectrum preamble.8. The multiple-access spread spectrum communication system of claim 7,wherein said spread spectrum preamble is the same for each of said userstations communicating with said base station.
 9. The multiple-accessspread spectrum communication system of claim 8, wherein said basestation is located in a cell within a cellular communication system,wherein said spread spectrum preamble comprises a spread spectrum codeselected from a set of spread spectrum codes available to each userstation, and wherein each user station transmits a second spreadspectrum preamble to a second base station located in a different cellfrom said base station, said second spread spectrum preamble comprisinga different spread spectrum code than said first spread spectrumpreamble.
 10. The multiple-access spread spectrum communication systemof claim 7, wherein said spread spectrum encoded data message comprisesa sequence of M-ary spread spectrum codes of predefined length, andwherein said base station receiver comprises means for dividing areceived signal into a real signal and an imaginary signal; anon-coherent parallel correlator for correlating to said spread spectrumpreamble, said non-coherent parallel correlator receiving as inputs saidreal signal and said imaginary signal and generating, during thepreamble sounding gap following receipt of said spread spectrumpreamble, a correlation signal of variable width upon recognizing saidspread spectrum preamble; a center-seeking circuit for identifying thecenter of correlation signal and synchronizing a correlation clockthereby during said preamble sounding gap; a bank of serial correlators,each of said serial correlators configured to recognize a different oneof said M-ary spread spectrum codes and generate a correlation signalupon recognizing its respective M-ary spread spectrum code, said serialcorrelators each comprising an integrator responsive to said correlationclock and each receiving as inputs said real signal and said imaginarysignal; and a best-of-M magnitude comparator and data extractorconnected to the output correlation signal from each of said serialcorrelators.
 11. A method for multiple-access spread spectrumcommunication, comprising the steps of: transmitting, from each of aplurality of user stations, a spread spectrum preamble; transmitting,from each of said user stations, a spread spectrum encoded data messagefollowing the respective spread spectrum preamble transmitted by a userstation, said spread spectrum encoded data message lagging said spreadspectrum preamble by a preamble sounding gap during which the userstation does not receive a data message, said preamble sounding gapshorter in duration than said spread spectrum preamble; receiving saidspread spectrum preambles at a receiver; generating a correlation signalat said receiver in response to each spread spectrum preamble receivedat said receiver; receiving said spread spectrum encoded data messagesat said receiver, each spread spectrum encoded data message receivedfollowing arrival of its respective spread spectrum preamble;despreading each spread spectrum encoded data message at said receiver.12. The method of claim 11, further comprising the steps of generating,a plurality of time frames, and defining a plurality of equally spacedtime offsets relative to a starting boundary of each of said timeframes, wherein said step of transmitting, from each of said pluralityof user stations, said spread spectrum preamble comprises the step oftransmitting, from each of said plurality of user stations, said spreadspectrum preamble with reference to one of said time offsets.